Code division multiple access wireless system with closed loop mode using ninety degree phase rotation and beamformer verification Number:7,394,867 from the United States Patent and Trademark Office (PTO) owispatent

Title: Code division multiple access wireless system with closed loop mode using ninety degree phase rotation and beamformer verification

Abstract: A wireless communication system (10). The system comprises a user station (12). The user station comprises despreading circuitry (22) for receiving and despreading a plurality of slots received from at least a first transmit antenna (A12.sub.1) and a second transmit antenna (A12.sub.2) at a transmitting station (14). Each of the plurality of slots comprises a first channel (DPCH) comprising a first set of pilot symbols and a second channel (PCCPCH) comprising a second set of pilot symbols. The user station further comprises circuitry (50) for measuring a first channel measurement (.alpha..sub.1,n) for each given slot in the plurality of slots from the first transmit antenna and in response to the first set of pilot symbols in the given slot The user station further comprises circuitry (50) for measuring a second channel measurement (.alpha..sub.2,n) for each given slot in the plurality of slots from the second transmit antenna and in response to the first set of pilot symbols in the given slot. The user station further comprises circuitry (52) for measuring a phase difference value (.phi..sub.2(n)) for each given slot in the plurality of slots in response to the first channel measurement and the second channel measurement for the given slot and in response to a ninety degree rotation of the given slot relative to a slot which was received by the despreading circuitry immediately preceding the given slot.

Patent Number: 7,394,867 Issued on 07/01/2008 to Dabak,   et al.

---

Inventors:	Dabak; Anand G. (Plano, TX), Onggosanusi; Eko N. (Madison, WI)
Assignee:	Texas Instruments Incorporated (Dallas, TX)
Appl. No.:	10/781,472
Filed:	February 17, 2004

---

Current U.S. Class:	375/295 ; 375/146
Current International Class:	H04L 27/00 (20060101); H04B 1/00 (20060101)
Field of Search:	375/358,267,299,347,141-144,146-151,295 455/101

---

References Cited [Referenced By]

---

U.S. Patent Documents

5848103	December 1998	Weerackody
5920286	July 1999	Mohebbi
6067324	May 2000	Harrison
6373433	April 2002	Espax et al.
6611675	August 2003	Salonen et al.

Other References

"3.sup.rd Generation Partnership Project (3GPP); Technical Specification Group (TSG) Radio Access Network (RAN) Working Group 1 (WG1); Physical Layer Procedures (FDD)", 3GPP Ran 25.214 V1, 10.01 (May 1999), TSGR1#5 (99)766, pp. 1-29, See Chapter 8. cited by other.

Primary Examiner: Bocure; Tesfaldet
Attorney, Agent or Firm: Neerings; Ronald O. Brady, III; Wade J. Telecky, Jr.; Frederick J.

---

Claims

---

That which is claimed:

1. A method of transmitting information comprising the steps of: receiving an information signal; receiving a plurality of coefficients from a remote communication system; averaging less than four of the coefficients over a plurality of slots; producing a plurality of weighted information signals from respective coefficients and the information signal; and transmitting the plurality of weighted information signals from respective antennas.

2. A method as in claim 1, comprising the steps of: encoding the information signal; interleaving the information signal; symbol mapping the information signal; and modulating the information signal.

3. A method as in claim 1, wherein the step of producing a plurality of weighted information signals comprises the steps of: multiplying the information signal by a first coefficient, thereby producing a first weighted information signal; and multiplying the information signal by a second coefficient, thereby producing a second weighted information signal.

4. A method as in claim 3 comprising the steps of: transmitting the first weighted information signal from a first antenna; and transmitting the second weighted information signal from a second antenna.

5. A method as in claim 3, wherein the respective coefficients correspond respectively to previously transmitted weighted information signals.

6. A method as in claim 3 comprising the steps of: transmitting a first set of pilot symbols over a primary common control physical channel (PCCPCH); and transmitting a second set of pilot symbols and the weighted information signals over a dedicated physical channel (DPCH).

7. The method as in claim 1, wherein less than four is two.

8. An apparatus, comprising: circuitry for receiving an information signal; circuitry for receiving a plurality of coefficients from a remote communication system; circuitry for averaging less than four of the coefficients over a plurality of slots; producing a plurality of weighted information signals from respective coefficients and the information signal; and transmitting the plurality of weighted information signals from respective antennas.

9. The apparatus of claim 8, further comprising: circuitry encoding the information signal; circuitry for interleaving the information signal; circuitry for symbol mapping the information signal; and circuitry for modulating the information signal.

10. The apparatus of claim 8, further comprising: circuitry for multiplying the information signal by a first coefficient, thereby producing a first weighted information signal; and circuitry for multiplying the information signal by a second coefficient, thereby producing a second weighted information signal.

11. The apparatus of claim 10, further comprising: circuitry for transmitting first weighted information signal from a first antenna; and circuitry for transmitting second weighted information signal from a second antenna.

12. The apparatus of claim 10, wherein the respective coefficients correspond respectively to previously transmitted weighted information signals.

13. The apparatus of claim 10, further comprising: circuitry for transmitting a first set of pilot symbols over a primary common control physical channel (PCCPCH); and circuitry for transmitting a second set of pilot symbols and the weighted information signals over a dedicated physical channel (DPCH).

14. The apparatus of claim 8, wherein less than four is two.

---

Description

---

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

This application claims the benefit, under 35 U.S.C. .sctn.119(e)(1), of U.S. Provisional Application No. 60/148,972 (TI-29547PS), filed Aug. 13, 1999, and incorporated herein by this reference.

CROSS-REFERENCES TO RELATED APPLICATIONS

Not Applicable.

BACKGROUND OF THE INVENTION

The present embodiments relate to wireless communications systems and, more particularly, to a closed-loop mode of operation for such systems.

Wireless communications have become very prevalent in business, personal, and other applications, and as a result the technology for such communications continues to advance in various areas. One such advancement includes the use of spread spectrum communications, including that of code division multiple access ("CDMA"). In such communications, a user station (e.g., a hand held cellular phone) communicates with a base station, where typically the base station corresponds to a "cell." Further, CDMA systems are characterized by simultaneous transmission of different data signals over a common channel by assigning each signal a unique code. This unique code is matched with a code of a selected user station within the cell to determine the proper recipient of a data signal. CDMA continues to advance and with such advancement there has brought forth a next generation wideband CDMA ("WCDMA"). WCDMA includes alternative methods of data transfer, one being frequency division duplex ("FDD") and another being time division duplex ("TDD").

Due to various factors including the fact that CDMA communications are along a wireless medium, an originally transmitted communication from a base station to a user station may arrive at the user station at multiple and different times. Each different arriving signal that is based on the same original communication is said to have a diversity with respect to other arriving signals originating from the same transmitted communication. Further, various diversity types may occur in CDMA communications, and the CDMA art strives to ultimately receive and identify the originally transmitted data by exploiting the effects on each signal that are caused by the one or more diversities affecting the signal.

One type of CDMA diversity occurs because a transmitted signal from a base station is reflected by objects such as the ground, mountains, buildings, and other things that it contacts. As a result, a same single transmitted communication may arrive at a receiving user station at numerous different times, and assuming that each such arrival is sufficiently separated in time, then each different arriving signal is said to travel along a different channel and arrive as a different "path." These multiple signals are referred to in the art as multiple paths or multipaths. Several multipaths may eventually arrive at the user station and the channel traveled by each may cause each path to have a different phase, amplitude, and signal-to-noise ratio ("SNR"). Accordingly, for one communication from one base station to one user station, each multipath is originally a replica of the same originally transmitted data, and each path is said to have time diversity relative to other multipath(s) due to the difference in arrival time which causes different (uncorrelated) fading/noise characteristics for each multipath. Although multipaths carry the same user data to the receiver, they may be separately recognized by the receiver based on the timing of arrival of each multipath. More particularly, CDMA communications are modulated using a spreading code which consists of a series of binary pulses, and this code runs at a higher rate than the symbol data rate and determines the actual transmission bandwidth. In the current industry, each piece of CDMA signal transmitted according to this code is said to be a "chip," where each chip corresponds to an element in the CDMA code. Thus, the chip frequency defines the rate of the CDMA code. Given the use of transmission of the CDMA signal using chips, then multipaths separated in time by more than one of these chips are distinguishable at the receiver because of the low auto-correlations of CDMA codes as known in the art.

In contrast to multipath diversity which is a natural phenomenon, other types of diversity are sometimes designed into CDMA systems in an effort to improve SNR, thereby improving other data accuracy measures (e.g., bit error rate ("BER"), frame error rate ("FER"), and symbol error rate ("SER")). An example of such a designed diversity scheme is antenna diversity and is introduced here since it pertains to the communication methodology used in the preferred embodiments discussed later. Looking first in general to antenna diversity, which is sometimes referred to as antenna array diversity, such diversity describes a wireless system using more than one antenna by a same station. Antenna diversity often proves useful because fading is independent across different antennas. Further, the notion of a station using multiple antennas is often associated with a base station using multiple antennas to receive signals transmitted from a single-antenna mobile user station, although more recently systems have been proposed for a base station using multiple antennas to transmit signals transmitted to a single-antenna mobile station. The present embodiments relate more readily to the case of a base station using multiple transmit antennas and, thus, this particular instance is further explored below.

The approach of using more than one transmit antenna at the base station is termed transmit antenna diversity. As an example in the field of mobile communications, a base station transmitter is equipped with two antennas for transmitting to a single-antenna mobile station. The use of multiple antennas at the base station for transmitting has been viewed as favorable over using multiple antennas at the mobile station because typically the mobile station is in the form of a hand-held or comparable device, and it is desirable for such a device to have lower power and processing requirements as compared to those at the base station. Thus, the reduced resources of the mobile station are less supportive of multiple antennas, whereas the relatively high-powered base station more readily lends itself to antenna diversity. In any event, transmit antenna diversity also provides a form of diversity from which SNR may be improved over single antenna communications by separately processing and combining the diverse signals for greater data accuracy at the receiver. Also in connection with transmit antenna diversity and to further contrast it with multipath diversity described above, note that the multiple transmit antennas at a single station are typically within several meters (e.g., three to four meters) of one another, and this spatial relationship is also sometimes referred to as providing spatial diversity. Given the spatial diversity distance, the same signal transmitted by each antenna will arrive at a destination (assuming no other diversity) at respective times that relate to the distance between the transmitting antennas. However, the difference between these times is considerably smaller than the width of a chip and, thus, the arriving signals are not separately distinguishable in the same manner as are multipaths described above.

Given the development of transmit antenna diversity schemes, two types of signal communication techniques have evolved to improve data recognition at the receiver given the transmit antenna diversity, namely, closed loop transmit diversity and open loop transmit diversity. Both closed loop transmit diversity and open loop transmit diversity have been implemented in various forms, but in all events the difference between the two schemes may be stated with respect to feedback. Specifically, a closed loop transmit diversity system includes a feedback communication channel while an open loop transmit diversity system does not. More particularly for the case of the closed loop transmit diversity system, a receiver receives a communication from a transmitter and then determines one or more values, or estimates, of the channel effect imposed on the received communication. The receiver then communicates (i.e., feeds back) one or more representations of the channel effect to the transmitter, so the transmitter may then modify future communication(s) in response to the channel effect. For purposes of the present document, the feedback values are referred to as beamformer coefficients in that they aid the transmitter in forming its communication "beam" to a user station.

With the advancement of CDMA and WCDMA there has been a comparable development of corresponding standards. For instance, a considerable standard that has developed, and which continues to evolve, in connection with WCDMA is the 3.sup.rd Generation partnership Project ("3GPP") for wireless communications, and it is also reflected in 3GPP 2 systems. Under 3GPP, closed loop antenna diversity for WCDMA must be supported, and in the past 3GPP set forth a closed loop operational method that alternates between three different communication modes. The choice of a mode at a given time is dictated by the Doppler fading rate of a particular user station receiver; in other words, since user stations are likely to be mobile, then due to the mobility as well as other factors there is likely to be an amount of Doppler fading in the signals received by such a user station from a base station and this fading affects the choice of a closed loop mode. In addition to the different fading rates giving rise to the selection of one of the three prior art modes of operation, each mode differs in certain respects. One difference is based on how the beamformer coefficients are quantized by the user station, and other differences also apply to different ones of the modes. Such differences are detailed later. In any event, note here by way of background that generally there is a tradeoff among the three modes, where greater resolution in the feedback information, and hence a greater level of beamformer control, is achieved at the expense of increased feedback and processing delay.

The preceding three modes have proven to achieve a considerable level of performance as measurable in various manners, such as BER, FER, or SNR; however, the present inventors also have identified various drawbacks with the overall three mode approach. For example, a certain level of complexity is required to implement the necessary algorithm to switch between the three different modes in response to changes in Doppler fading. As another example, an alternative approach may be implemented using one mode which provides results that match or outperform the results achieved by the prior art modes 1 and 2 across the Doppler frequencies for which those prior art modes are used. Still other benefits may be ascertainable by one skilled in the art given a further understanding of the preferred embodiments, as should be accomplished from the detailed description provided below.

BRIEF SUMMARY OF THE INVENTION

In the preferred embodiment, there is a wireless communication system. The system comprises a user station. The user station comprises despreading circuitry for receiving and despreading a plurality of slots received from at least a first transmit antenna and a second transmit antenna at a transmitting station. Each of the plurality of slots comprises a first channel comprising a first set of pilot symbols and a second channel comprising a second set of pilot symbols. The user station further comprises circuitry for measuring a first channel measurement for each given slot in the plurality of slots from the first transmit antenna and in response to the first set of pilot symbols in the given slot. The user station further comprises circuitry for measuring a second channel measurement for each given slot in the plurality of slots from the second transmit antenna and in response to the first set of pilot symbols in the given slot. The user station further comprises circuitry for measuring a phase difference value for each given slot in the plurality of slots in response to the first channel measurement and the second channel measurement for the given slot and in response to a ninety degree rotation of the given slot relative to a slot which was received by the despreading circuitry immediately preceding the given slot.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

FIG. 1 illustrates a closed loop transmit antenna diversity system within which the preferred embodiments may be implemented.

FIG. 2 illustrates an expanded view of selected blocks of user station 14 from FIG. 1.

FIG. 3 illustrates a graph to depict the prior art mode 1 mapping of a channel measurement to one of two different phase shift values.

FIG. 4 illustrates four graphs to depict the prior art mode 2 mapping of channel measurements according to respective 45 degree rotations, where for each rotation the channel measurement is mapped to one of two different phase shift values.

FIG. 5 illustrates two graphs to depict the mapping of channel measurements according to the preferred embodiment broad range closed loop mode according to respective 90 degree rotations, where for each rotation the channel measurement is mapped to one of two different phase shift values.

FIG. 6 illustrates a block diagram of the functional operation of beamformer coefficient computation block 52 and beamformer coefficient binary encode block 54 from FIG. 2 and according to the preferred embodiment.

FIG. 7 illustrates a block diagram of channel estimation and beamformer verification block 56 from FIG. 2 and according to the preferred embodiment.

FIG. 8 illustrates a block diagram of a first implementation of a beamformer verification block 100.sub.1 that may readily implemented as beamformer verification block 100 from FIG. 7, and which operates according to a two rotating hypothesis testing method.

FIG. 9 illustrates a block diagram of a second implementation of a beamformer verification block 100.sub.2 that also may be implemented as beamformer verification block 100 from FIG. 7, and which operates according to a four hypothesis single shot testing.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates a closed loop transmit antenna diversity system 10 within which the preferred embodiments may be implemented, and also which in a block form may represent the prior art. Accordingly, the following discussion first examines system 10 in a general fashion as applying to both the preferred embodiments and the prior art, followed by a detailed discussion with additional illustrations of the particular modifications to system 10 to implement the preferred embodiments.

Turning to system 10 of FIG. 1, it includes a transmitter 12 and a receiver 14. By way of example, assume that transmitter 12 is a base station 12 while receiver 14 is a mobile user station 14. Also, for the sake of simplifying the discussion, each of these components is discussed separately below. Lastly, note that the closed loop technique implemented by system 10 is sometimes referred to in the art as a transmit adaptive array ("TxAA"), while other closed loop techniques also should be ascertainable by one skilled in the art.

Base station 12 receives information bits B.sub.i at an input to a channel encoder 13. Channel encoder 13 encodes the information bits B.sub.i in an effort to improve raw bit error rate. Various encoding techniques may be used by channel encoder 13 and as applied to bits B.sub.i, with examples including the use of convolutional code, block code, turbo code, concatenated codes, or a combination of any of these codes. The encoded output of channel encoder 13 is coupled to the input of an interleaver 15. Interleaver 15 operates with respect to a block of encoded bits and shuffles the ordering of those bits so that the combination of this operation with the encoding by channel encoder 13 exploits the time diversity of the information. For example, one shuffling technique that may be performed by interleaver 15 is to receive bits in a matrix fashion such that bits are received into a matrix in a row-by-row fashion, and then those bits are output from the matrix to a symbol mapper 16 in a column-by-column fashion. Symbol mapper 16 then converts its input bits to symbols, designated generally as S.sub.i. The converted symbols S.sub.i may take various forms, such as quadrature phase shift keying ("QPSK") symbols, binary phase shift keying ("BPSK") symbols, or quadrature amplitude modulation ("QAM") symbols. In any event, symbols 5, may represent various information such as user data symbols, as well as pilot symbols and control symbols such as transmit power control ("TPC") symbols and rate information ("RI") symbols. Symbols S.sub.i are coupled to a modulator 18. Modulator 18 modulates each data symbol by combining it with, or multiplying it times, a CDMA spreading sequence which can be a pseudo-noise ("PN") digital signal or PN code or other spreading codes (i.e., it utilizes spread spectrum technology). In any event, the spreading sequence facilitates simultaneous transmission of information over a common channel by assigning each of the transmitted signals a unique code during transmission. Further, this unique code makes the simultaneously transmitted signals over the same bandwidth distinguishable at user station 14 (or other receivers). Modulator 18 has two outputs, a first output 18.sub.1 connected to a multiplier 20.sub.1 and a second output 18.sub.2 connected to a multiplier 20.sub.2. Generally, each of multipliers 20.sub.1 and 20.sub.2, for a communication slot n, receives a corresponding and per-slot decoded weight value, .omega..sub.1,T(n) and .omega..sub.2,T(n), from a feedback decode and process block 21. Feedback decode and process block 21 provides weighted values, .omega..sub.1,T(n) and .omega..sub.2,T(n) in response to values .omega..sub.1(n) and .omega..sub.2(n), respectively, as further discussed below. Each of multiplier 20.sub.1 and 20.sub.2 multiplies the respective value .omega..sub.1,T(n) and .omega..sub.2,T(n) times the corresponding output 18.sub.1 or 18.sub.2 from modulator 18 and, in response, each of multipliers 20.sub.1 and 20.sub.2 provides an output to a respective transmit antenna A12.sub.1 and A12.sub.2, where antennas A12.sub.1 and A12.sub.2 are approximately three to four meters apart from one another. As detailed later, in applying the various modes of operation in the prior art, the operation of multiplier 20.sub.1 is based on normalized value (i.e., .omega..sub.1,T(n) is normalized), while the operation of multiplier 20.sub.2 may be based on a single slot value of .omega..sub.2,T(n) for certain modes of operation while it is based on an average of successively received values of .omega..sub.2,T(n) for another mode of operation, and in either case .omega..sub.2,T(n) is relative to the normalized value of .omega..sub.1,T(n).

Receiver 14 includes a receive antenna A14.sub.1 for receiving communications from both of transmit antennas A12.sub.1 and A12.sub.2. Recall that such communications may pass by various multipaths, and due to the spatial relationship of transmit antennas A12.sub.1 and A12.sub.2, each multipath may include a communication from both transmit antenna A12.sub.1 and transmit antenna A12.sub.2. In the illustration of FIG. 1, a total of P multipaths are shown. Within receiver 14, signals received by antenna A14.sub.1 are connected to a despreader 22. Despreader 22 operates in many respects according to known principles, such as by multiplying the CDMA signal times the CDMA code for user station 14 and resolving any multipaths, thereby producing a despread symbol stream at its output and at the symbol rate. Additional details relating to despreader 22 are also discussed later in connection with its breakdown of different channels of information as received by antenna A14.sub.1. The despread signals output by despreader 22 are coupled to maximal ratio combining ("MRC") block 23, and also to a channel evaluator 24. As detailed considerably below, channel evaluator 24 performs two different channel determinations, and to avoid confusion one such determination is referred to as channel measurement while the other is referred to as channel estimation, where both determinations are based at least on the incoming despread data. Further, channel evaluator 24 provides two outputs. A first output 24.sub.1 from channel estimator 24 outputs a channel estimation, designated as {acute over (h)}.sub.n, to MRC block 23. In response to receiving the channel estimation, MRC block 23 applies the estimation to the despread data symbols received from despreader 22 using a rake receiver; however, the application of the estimate to the data may be by way of alternative signal combining methods. A second output 24.sub.2 from channel evaluator 24 communicates the values .omega..sub.1(n) and .omega..sub.2(n), introduced earlier, back toward base station 12 via a feedback channel. As also detailed below, the values .omega..sub.1(n) and .omega..sub.2(n) are determined by channel evaluator 24 in response to a channel measurement made by channel evaluator 24. In any event, one skilled in the art should appreciate from the preceding that the values .omega..sub.1(n) and .omega..sub.2(n) are therefore the closed loop beamformer coefficients introduced above.

Returning to MRC block 23 of user station 14, once it applies the channel estimation to the despread data, its result is output to a deinterleaver 25 which operates to perform an inverse of the function of interleaver 15, and the output of deinterleaver 25 is connected to a channel decoder 26. Channel decoder 26 may include a Viterbi decoder, a turbo decoder, a block decoder (e.g., Reed-Solomon decoding), or still other appropriate decoding schemes as known in the art. In any event, channel decoder 26 further decodes the data received at its input, typically operating with respect to certain error correcting codes, and it outputs a resulting stream of decoded symbols. Indeed, note that the probability of error for data input to channel decoder 26 is far greater than that after processing and output by channel decoder 26. For example, under current standards, the probability of error in the output of channel decoder 26 may be between 10.sup.-3 and 10.sup.-6. Finally, the decoded symbol stream output by channel decoder 26 may be received and processed by additional circuitry in user station 14, although such circuitry is not shown in FIG. 1 so as to simplify the present illustration and discussion.

Having detailed system 10, attention is now returned to its identification as a closed loop system. Specifically, system 10 is named a closed loop system because, in addition to the data communication channels from base station 12 to user station 14, system 10 includes the feedback communication channel for communicating the beamformer coefficients .omega..sub.1(n) and .omega..sub.2(n) from user station 14 to base station 12; thus, the data communication and feedback communication channels create a circular and, hence, "closed" loop system. Note further that beamformer coefficients .omega..sub.1(n) and .omega..sub.2(n) may reflect various channel affecting aspects. For example, user station 14 may ascertain a level of fading in signals it receives from base station 12, such as may be caused by local interference and other causes such as the Doppler rate of user station 14 (as a mobile station), and in any event where the fading may be characterized by Rayleigh fading. As a result, user station 14 feeds back beamformer coefficients .omega..sub.1(n) and .omega..sub.2(n), and these values are processed by feedback decode and process block 21 to produce corresponding values .omega..sub.1,T(n) and .omega..sub.2,T(n), which are used by multipliers 20.sub.1 and 20.sub.2 to apply those values to various symbols to provide respective resulting transmitted signals along transmitter antenna A12.sub.1 (in response to .omega..sub.1,T(n)) and along transmitter antenna A12.sub.2 (in response to .omega..sub.2,T(n)). Thus, for a first symbol S.sub.1 to be transmitted by base station 12, it is transmitted as part of a product .omega..sub.1,T(n)S.sub.1 along transmitter antenna A12.sub.1 and also as part of a product .omega..sub.2,T(n)S.sub.1 along transmitter antenna A12.sub.2. By way of illustration, therefore, these weighted products are also shown in FIG. 1 along their respective antennas.

Having detailed closed loop transmit antenna diversity systems, attention is now directed to the above-introduced 3GPP standard and its choice of closed loop modes at a given time in response to the Doppler fading rate of a particular user station receiver. Specifically, the following Table 1 illustrates the three different former 3GPP closed diversity modes and correlates each mode to an approximate Doppler fading rate (i.e., frequency).

TABLE-US-00001 TABLE 1 Prior Art Mode Doppler fading rate, f(Hz) 1 f > 60 2 10 < f < 60 3 f < 10

In addition to the different fading rates giving rise to the selection of one the three prior art modes of operation in Table 1, the methodology of each mode differs in certain respects. One difference is based on how the beamformer coefficients (e.g., .omega..sub.1(n) and .omega..sub.2(n) from FIG. 1) are quantized, and other differences also apply to different ones of the modes. Such differences are further explored below.

Looking to the prior art mode 1 of operation from Table 1, it is used for relatively high Doppler fading rates, such as would be expected when the particular mobile user station 14 with which base station 12 is communicating is moving at a relatively large rate of speed. To accommodate the higher Doppler fade, mode 1 uses a reduced amount of quantization for the beamformer coefficients, that is, in mode 1 the user station feeds back a lesser amount of information to represent these coefficients. More particularly, in mode 1, a beamformer coefficient vector W is fed back by the user station, and for a two antenna base station let that coefficient vector be represented in the following Equation 1: W=(.omega..sub.1(n), .omega..sub.2(n)) Equation 1 In Equation 1, the coefficient .omega..sub.1(n) is intended to apply to base station transmit antenna A12.sub.1 while the coefficient .omega..sub.2(n) is intended to apply to base station transmit antenna A12.sub.2. In practice and to further reduce the amount of feedback information, .omega..sub.1(n) is normalized to a fixed value and, thus, it is not necessary to feed it back so long as the normalized value is known by base station 12. Accordingly, when .omega..sub.1(n) is normalized, only the value of .omega..sub.2(n) may change and is relative to the fixed value of .omega..sub.1(n) and, therefore, .omega..sub.2(n) is fed back from user station 14 to base station 12. Further, in the prior art mode 1, .omega..sub.2(n) is only allowed to be one of two values. The quantizations offered by the vector W therefore may be represented by the following Equations 2 and 3: W=(1, 0) Equation 2 W=(1, 1) Equation 3 Thus, mode 1 only requires the feedback of one of two values (i.e., for .omega..sub.2(n)). Further, note that the conventions in Equations 2 and 3 depict binary values, while one skilled in the art should appreciate that for the case of a binary 0, an actual value of -1 is provided on the physical feedback channel, while for the case of a binary 1, an actual value of +1 is provided on the physical feedback channel. Finally, the prior art manner for selecting the value of .omega..sub.2(n), that is, between binary 0 and 1, is discussed below.

The prior art mode 1 determination of .omega..sub.2(n) is better appreciated from the expanded illustration of FIG. 2, where certain blocks of user station 14 from FIG. 1 are further detailed. Looking to FIG. 2, it again illustrates antenna A14.sub.1 providing signals to despreader 22. Despreader 22 is expanded in FIG. 2 to illustrate that it includes a despreading and resolve multipath block 40. Block 40 despreads the incoming signals from two different channels, that is, recall it was earlier introduced that despreader 22 operates with respect to different channels of information as received by antenna A14.sub.1; these different channels are now illustrated in FIG. 2 as a primary common control physical channel ("PCCPCH") and a dedicated physical channel ("DPCH"). According to the prior art, the PCCPCH is transmitted by base station 12 as the same channel to all user stations (i.e., user station 14 and others communicating with base station 12), and it is not weighted in response to .omega..sub.1(n) and .omega..sub.2(n). The DPCH, however, is user station specific and it is weighted in response to .omega..sub.1(n) and .omega..sub.2(n). Both the PCCPCH and DPCH communicate in frame formats, where each frame includes a number of slots; for example, in WCDMA, each frame consists of 16 slots. Further with respect to PCCPCH and DPCH, each slot of those channels commences with some pilot symbols and also includes information symbols. Given the preceding, block 40 operates with respect to each received slot and outputs both a DPCH symbol stream and PCCPCH symbol stream, and the further processing of those streams is discussed below.

The DPCH symbol stream from block 40 is coupled to both an information symbol extractor 42 and a pilot symbol extractor 44. Each of blocks 42 and 44 operates as suggested by their names, that is, to extract from the DPCH symbol stream the DPCH information symbols and the DPCH pilot symbols, respectively. For sake of reference in this document, the DPCH information symbols are represented by x(n) while the DPCH pilot symbols are represented by y(n), where bold face is used as a convention for these and other values in this document to indicate that the value is a vector. The DPCH information symbols x(n) are output from extractor 42 to MRC block 23, while the DPCH pilot symbols y(n) are output from extractor 44 to channel evaluator 24, as further detailed later.

Returning to despreading and resolve multipath block 40 and its output of the PCCPCH symbol stream, that stream is coupled to a PCCPCH pilot symbol extractor 46. PCCPCH pilot symbol extractor 46 extracts the PCCPCH pilot symbols from the PCCPCH symbol stream. For sake of reference in this document, the PCCPCH pilot symbols are represented by z(n). The PCCPCH pilot symbols z(n) are output from extractor 46 to channel evaluator 24, as further detailed below.

Looking to channel evaluator 24 in FIG. 2, it includes a channel measurement block 50 which receives the PCCPCH pilot symbols z(n) from extractor 46. Recalling that it was earlier stated that channel evaluator 24 performs both a channel measurement and channel estimation based at least on the incoming despread data, it is now noted more particularly that block 50 performs the channel measurement aspect. Specifically, the PCCPCH pilot symbols are, according to the art, different for each different transmit antenna for a base station; thus, in the present example, the extracted PCCPCH pilot symbols z(n) includes one set of pilot symbols corresponding to base station antenna A12.sub.1 and another set of pilot symbols corresponding to base station antenna A12.sub.2. Since the values of the pilot symbols as transmitted by base station 12 are by definition a known value to user station 14, then based on the difference between the actually received pilot symbols and the known transmitted pilot symbols, block 50 determines, for each transmit antenna, a channel measurement reflecting any change in the actually-received pilot symbols. For sake of reference in this document, the channel measurement corresponding to antenna A12.sub.1 is indicated as .alpha..sub.1,n and the channel measurement corresponding to antenna A12.sub.2 is indicated as .alpha..sub.2,n. Both .alpha..sub.1,n and .alpha..sub.2,n are output by channel measurement block 50 to a beamformer coefficient computation block 52.

Beamformer coefficient computation block 52 computes phase difference values, denoted .phi..sub.1(n) and .phi..sub.2(n), in response to the values .alpha..sub.1,n and .alpha..sub.2,n, where the values .phi..sub.1(n) and .phi..sub.2(n) as described below are the angular phase differences which are encoded into binary form to create the respective values of .omega..sub.1(n) and .omega..sub.2(n) (or just .omega..sub.2(n) where .omega..sub.1(n) is a normalized value). Recall now that under mode 1 of the prior art, the value of .omega..sub.2(n) may be only one of two states. Thus, block 52 maps the value of .alpha..sub.2,n to one of these two states, and this mapping function is illustrated pictorially in FIG. 3 by a graph 52g plotted along an imaginary and real axis. More particularly, graph 52g illustrates two shaded areas 52.sub.1 and 52.sub.2 corresponding to the two possible values of .omega..sub.2(n), and those two values map to two corresponding values of .phi..sub.2(n). Specifically, if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 52.sub.1, then the value of .phi..sub.2(n) is 0 degrees; further, this 0 degree value of .phi..sub.2(n) is output to a beamformer coefficient binary encode block 54 which converts the angular value .phi..sub.2(n) of 0 degrees into a corresponding binary value .omega..sub.2(n)=0, and the value of .omega..sub.2(n)=0 is fed back to base station 12. On the other hand, if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 52.sub.2 then the value of .phi..sub.2(n) is .pi. degrees; further, this .pi. degree value of .phi..sub.2(n) is output to beamformer coefficient binary encode block 54 which converts the angular value .phi..sub.2(n) of .pi. degrees into a corresponding binary value .omega..sub.2(n)=1, and the value of .omega..sub.2(n)=1 is fed back to base station 12.

Attention is now directed to an additional aspect of the prior art mode 1. Specifically, note that while user station 14 transmits a value of .omega..sub.2(n) to base station 12, there quite clearly can be effects imposed on that transmission as well, that is, there is a channel effect in the feedback signal from user station 14 to base station 12. Accordingly, from the perspective of base station 12, let {tilde over (.omega.)}.sub.2(n) represent the signal actually received by base station 12 and corresponding to the feedback transmission of .omega..sub.2(n) from user station 14. Next, feedback decode and process block 21 decodes and processes {tilde over (.omega.)}.sub.2(n) and in response outputs a corresponding value of .omega..sub.2,T(N) which is multipled by multiplier 18.sub.2. As a result, while ideally base station 12 uses the correct value .omega..sub.2(n) upon which to determine .omega..sub.2,T(n) and to create a resulting product signal (i.e., .omega..sub.2,T(n)S.sub.i)), the feedback channel effect may cause base station 12 to use a different value of .omega..sub.2(n). For example, user station 14 may transmit a value of .omega..sub.2(n)=0 to base station 12, but due to the feedback channel the received value may be {tilde over (.omega.)}.sub.2(n)=1. Conversely, user station 14 may transmit a value of .omega..sub.2(n)=1 to base station 12, but due to the feedback channel the received value may be {tilde over (.omega.)}.sub.2(n)=0. In view of these possibilities, user station 14, when operating under the prior art mode 1, further implements a process referred to in the art as beamformer verification or antenna verification, as further detailed below.

Beamformer verification is further introduced by return to the expanded block diagram in FIG. 2. Specifically, recall it is stated above that the DPCH pilot symbols y(n) are output from extractor 44 to channel evaluator 24, and recall also that the DPCH pilot symbols have been modified by base station 12 in response to .omega..sub.1,T(n) and .omega..sub.2,T(n). Further, and as now discussed and as shown in FIG. 2, the DPCH pilot symbols y(n) are connected to a channel estimation and beamformer verification block 56. Block 56 also receives as inputs the channel measurement values .alpha..sub.1,n and .alpha..sub.2,n from channel measurement block 50 and the phase difference values .phi..sub.1(n) and .phi..sub.2(n) from beamformer coefficient computation block 52. In response to its inputs, block 56 outputs the channel estimation, introduced earlier as h.sub.n, to MRC block 23, but in doing so the beamformer verification process attempts to ensure that h.sub.n is correctly estimated in view of previously fed back beamformer coefficients. Specifically, note that h.sub.n may be defined according to the following Equation 4: h.sub.n=.alpha..sub.1,n.omega..sub.1,T(n)+.alpha..sub.2,n.omega..sub.2,T(- n) Equation 4 Thus, Equation 4 indicates mathematically that the overall change (i.e., the channel estimation, h.sub.n) in a signal received by user station 14 should be reflected by both the channel measurement factors .alpha..sub.1,n and a.sub.2,n as well as the weight factors .omega..sub.1,T(n) and .omega..sub.2,T(n) that were multiplied by base station 12 against the signal before it was transmitted by base station 12 to user station 14. Thus, beamformer verification is a process by which user station 14 attempts to ascertain .omega..sub.1,T(n) and .omega..sub.2,T(n) as used by base station 12, and those values may then be used to determine h.sub.n.

Equation 4 also demonstrates that, in one approach, the channel estimation, h.sub.n, could be a direct calculation because block 56 receives the channel measurement values .alpha..sub.1,n and .alpha..sub.2,n and if it is assumed that .omega..sub.1,T(n) and .omega..sub.2,T(n) could be identified from the phase difference values .phi..sub.1(n) and .phi..sub.2(n) from beamformer coefficient computation block 52. However, because base station 12 responds to {tilde over (.omega.)}.sub.2(n) rather than .omega..sub.2(n), then beamformer verification is a process by which user station 14 attempts to predict what value of {tilde over (.omega.)}.sub.2 (n) was received by base station 12 and that predicted value may then be used to identify the counterpart .omega..sub.2,t(n) in Equation 4 to determine h.sub.n. To further appreciate this concept, beamformer verification also may be understood in connection with an example. Thus, suppose for a slot n=1, user station 14 transmits a feedback value of .omega..sub.2(1) to base station 12; in response, base station 12 receives a value, {tilde over (.omega.)}.sub.2(1), block 21 produces a corresponding value .omega..sub.2,T(1), and a product .omega..sub.2,T(1)S.sub.i is formed and transmitted next to user station 14. Under beamformer verification as used in the prior art mode 1, user station 14 receives the signal .omega..sub.2,T(1)S.sub.i, and from that signal it attempts to determine what value of .omega..sub.2,T(1) was actually used by base station 12 in its corresponding transmission, and this attempt is achieved by block 56 using a methodology referred to as hypothesis testing. This determined value, rather than the actual value .omega..sub.2(1) which was fed back by user station 14, is then used by block 56 to determine h.sub.n, and that value of h.sub.n is used by MRC block 23 for further signal processing.

Concluding the discussion of the prior art mode 1, note that its use of only two possible data values for .omega..sub.2(n), in combination with the operations relating to hypothesis testing, have yielded a workable error rate at a reasonable level of performance speed. Indeed, relative to prior art modes 2 and 3 described below, the feedback delay of prior art mode 1 is relatively small, and a certain level of performance is achieved given this reduced delay. However, the resulting resolution obtained in response to the 2-state level of quantization of mode 1 is relatively low as compared to prior art modes 2 and 3 as further discussed below.

Looking to the prior art mode 2 of operation from Table 1, it is used for relatively mid-level Doppler fading rates, such as would be expected when a particular mobile user station 14 with which base station 12 is communicating is moving at a lesser rate of speed than for the case of a mode 1 communication. Mode 2 again uses the convention of Equation 1 and normalizes .omega..sub.1(n) (and thereby its counterpart .phi..sub.1(n)), but added resolution is obtained in the computation of .phi..sub.2(n) and .omega..sub.2(n) by beamformer coefficient computation block 52. Specifically, block 52 in mode 2 applies a 45 degree constellation rotation per slot to the 2-value beamformer coefficient, that is, for each successive slot, .phi..sub.2(n) and .omega..sub.2(n) are determined based on a 45 degree rotation relative to the preceding slot; particularly, since a total of four such rotations corresponds to 180 degrees, then under the 45 degree constellation rotation the slots are generally analyzed by user station 14 by adding a 45 degree rotation to each successive slot in each succession of four slots. This rotation is achieved at user station 14 by determining the value of .phi..sub.2(n) and .omega..sub.2(n) in part based on the time slot in which the slot at issue was received and then choosing the value of .omega..sub.2(n) with respect to the rotation to be applied to the given slot for a group i of four slots. This rotation is explored immediately below in connection with the following Table 2 and is also depicted pictorially in FIG. 4.

TABLE-US-00002 TABLE 2 w.sub.2(n) slot 4i slot 4i + 1 slot 4i + 2 slot 4i + 3 0 0 .pi./4 .pi./2 3.pi./4 1 .pi. -3.pi./4 -.pi./2 -.pi./4

Looking to Table 2 and FIG. 4, for a first slot 4i in a group i of four slots, the values of .phi..sub.2(n) and .omega..sub.2(n), as determined by beamformer coefficient computation block 52 in user station 14, are based on a rotation of zero degrees, as shown in graph 60 and as represented by its axis 60.sub.ax which is not rotated relative to the vertical imaginary axis. More particularly, graph 60 illustrates two shaded areas 60.sub.1 and 60.sub.2, where if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n from channel measurement block 50 falls within area 60.sub.1, then block 52 computes the value of .phi..sub.2(4i) to be 0 degrees and this value is encoded to a corresponding binary form .omega..sub.2(4i)=0 by encode block 54 and is fed back to base station 12; conversely, if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 60.sub.2, then block 52 computes the value of .phi..sub.2(4i) to be .pi. degrees and this value is encoded to a corresponding binary form .omega..sub.2(4i)=1 by block 54 and is fed back to base station 12. Further, Table 2 as well as the location of points on graph 60 illustrate the phase rotation that is implemented by base station 12 in response to the values of {tilde over (.omega.)}.sub.2(4i). Specifically, if base station 12 receives a value of {tilde over (.omega.)}.sub.2 (4i) equal to 0, then feedback decode and process block of base station 12 treats the channel measurement phase change for slot 4i as 0 degrees; however, if base station 12 receives a value of {tilde over (.omega.)}.sub.2(4i) equal to 1, then base station 12 treats the channel measurement phase change for slot 4i as equal to it degrees.

Table 2 and FIG. 4 also illustrate the remaining three slots in group i, where comparable reference numbers are used in FIG. 4 such that graph 62 corresponds to slot 4i+1 and represents a rotation equal to .pi./4 degrees, graph 64 corresponds to slot 4i+2 and represents a rotation equal to .pi./2 degrees, and graph 66 corresponds to slot 4i+3 and represents a rotation equal to 3.pi./4 degrees. Thus, looking to graph 62 as another example, its axis 62.sub.ax depicts the rotation of .pi./4 degrees relative to the vertical imaginary axis as used for slot 4i+1. Further, graph 62 illustrates two shaded areas 62.sub.1 and 62.sub.2, where if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n determined by block 50 of user station 14 falls within area 62.sub.1, then the value of .phi..sub.2(4i+1) is .pi./4 degrees and a corresponding binary value for .omega..sub.2(4i+1) equal to 0 is produced and fed back to base station 12, whereas if the channel measurement a .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 62.sub.2, then the value of .phi..sub.2(4i+1) is -3.pi./4 degrees and a corresponding binary value for .omega..sub.2(4i+1) equal to 1 is produced and fed back to base station 12. Further, Table 2 as well as the location of points on graph 62 illustrate the phase rotation that is implemented by base station 12 in response to the values of {tilde over (.omega.)}.sub.2(4i+1). Specifically for slot 4i+1, if base station 12 receives a value of {tilde over (w)}.sub.2 (4i+1) equal to 0, then feedback decode and process block of base station 12 treats the channel estimation phase change for slot 4i+1 as .pi./4 degrees; however, if base station 12 receives a value of {tilde over (.omega.)}.sub.2(4i+1) equal to 1, then base station 12 treats the channel estimation phase change for slot 4i+1 as equal to -3.pi./4 degrees. Given this second example as well as the preceding example described above, one skilled in the art should readily appreciate the remaining values and illustrations in Table 2 and FIG. 4 as applied to the value of .phi..sub.2(n) by user station 14 and the conversion of that value to .omega..sub.2(n) as well as the interpretation of the value of {tilde over (.omega.)}.sub.2v(n) by feedback decode and process block of base station 12 according to mode 2 in the prior art.

Attention is now directed to an additional aspect of the prior art mode 2 processing in response to .omega..sub.2(n) transmitted by user station 14. First, recalling the convention introduced above with respect to mode 1, from the perspective of base station 12, {tilde over (.omega.)}.sub.2 (n) represents the signal actually received by base station 12 and corresponding to the feedback transmission of .omega..sub.2(n) from user station 14. Second, note now that feedback decode and process block 21 during the prior art mode 2 actually uses an averaging filter to determine the value of .omega..sub.2,T(n) for each received value of {tilde over (.omega.)}.sub.2(n). Specifically, block 21 calculates an average over four values of .omega..sub.2 (or {tilde over (.omega.)}.sub.2, from the perspective of base station 12), so that the result is .omega..sub.2,T(n) and is defined by the following Equation 5:

.function..function..times..times..function..times..function..times..funct- ion..times..times..times. ##EQU00001## The indication of {tilde over (.omega.)}.sub.2(4i) in Equation 5 is to depict the most recent beamformer coefficient received by base station 12 via the feedback channel, and thus the remaining three addends in Equation 5 are based on the three other beamformer coefficients preceding that most recent coefficient. These four values are averaged (i.e., divided by 4), and in the prior art mode 2 of operation, base station 12 multiplies the result, .omega..sub.2,T(n), times the signal from second output 18.sub.2 connected to multiplier 20.sub.2. .omega..sub.1,T(n), however, is simply the counterpart to the normalized value .omega..sub.1(n), and base station 12 multiplies it times the signal from first output 18.sub.1, connected to multiplier 20.sub.1.

Given the preceding, one skilled in the art will appreciate that the prior art mode 2 also implements the feedback of one of two values (i.e., for .omega..sub.2(n)). However, given the additional use of phase rotation, greater beamformer resolution is achieved relative to prior art mode 1. In other words, while .omega..sub.2(n) for any given slot may only take one of two values as in the case of the prior art mode 1, the use of 45 degree rotation over four slots creates an effective constellation of eight possible values (i.e., 2 values/slot*4 slots/rotation cycle=8 values). However, note that the prior art mode 2 does not use any type of beamformer verification which is used by prior art mode 1; indeed, the present inventors have observed that beamformer verification may not be feasible for the prior art mode 2 because it could add, in combination with the four-cycle 45 degree rotation, an unworkable amount of complexity. Further, with phase rotation and averaging, the better resolution of the prior art mode 2 is offset in part by an increased overall delay relative to prior art mode 1.

Looking to the prior art mode 3 of operation from Table 1, it is appreciated as used for Doppler fading rates that are relatively low as compared to prior art modes 1 and 2, where the mode 3 fading rates would be expected when mobile user station 14 is moving at a relatively low rate of speed. Given the lower speed of user station 14, additional time is available for additional levels of processing, as is implemented in the prior art mode 3. Specifically, mode 3 increases its quantization for the beamformer coefficients, but the increase is not achieved based on rotation as shown in FIG. 2 for the prior art mode 2. Instead, the prior art mode 3 feeds back a total of four bits of information, where one bit is intended as an amplitude correction bit while the remaining three bits are to correct for phase shifts.

Having discussed closed loop transmit antenna diversity system 10 as it may implement the prior art, the attention is now directed to an implementation of the preferred embodiment into system 10. By way of overview to the preferred embodiment, it contemplates various alternative aspects versus those discussed above. First, in the preferred embodiment, prior art modes 1 and 2 are eliminated and replaced by a single mode of operation; because this single mode of operation spans the entire Doppler fading range of prior art modes 1 and 2, it is hereafter referred to as the broad range closed loop mode. Thus, the broad range closed loop mode may be combined with the prior art mode 3 of operation to accommodate the entire anticipated range of Doppler frequencies for closed loop communications. Second, with respect to the broad range closed loop mode, in addition to providing one mode in place of two prior art modes, it includes additional aspects that distinguish it further from the prior art. One such aspect is the use of a two phase rotation for determining beamformer coefficients. Another aspect is the use of beamformer verification, implemented using one of two different alternatives, in the same mode that implements phase rotation for determining beamformer coefficients. Each of these points should be further appreciated by one skilled in the art given the remaining teachings of this document.

The use of a two phase rotation for determining beamformer coefficients according to the broad range closed loop mode is now described. The broad range closed loop mode uses the earlier convention from Equation 1 and normalizes the value .omega..sub.1(n) as detailed later, and also thereby normalizes its phase difference counterpart, .phi..sub.1(n); however, in the preferred embodiment an overall resolution differing from the prior art modes 1 and 2 is obtained in the computation of .phi..sub.2(n) and .omega..sub.2(n) by beamformer coefficient computation block 52. Specifically, block 52 in the broad range closed loop mode applies a 90 degree constellation rotation per slot to the 2-value beamformer coefficient. Accordingly, for each successive slot n, n+1, n+2, and so forth, .phi..sub.2(n) and .omega..sub.2(n) are determined based on a 90 degree rotation relative to the preceding slot. Since a total of two such rotations correspond to 180 degrees, then under the 90 degree constellation rotation the slots are generally analyzed by user station 14 by adding a 90 degree rotation to each successive slot. This rotation is achieved at user station 14 by determining the value of .phi..sub.2(n) in part based on the time slot in which the slot at issue was received and then choosing the value of .phi..sub.2(n) with respect to the rotation to be applied to the given slot for a group i of two slots. This rotation is explored immediately below in connection with the following Table 3 and is also depicted pictorially in FIG. 5, and recall also that these operations may be implemented within system 10 shown in FIG. 1 to thereby create the preferred embodiment.

TABLE-US-00003 TABLE 3 w.sub.2(n) slot 2i slot 2i + 1 0 0 .pi./2 1 .pi. -.pi./2

Looking to Table 3 and FIG. 5, for a first slot 2i in a group i of two slots, the value of .phi..sub.2(2i), as determined by beamformer coefficient computation block 52 in user station 14, is based on a rotation of zero degrees as shown in graph 70 and as represented by its axis 70.sub.ax which is not rotated relative to the vertical imaginary axis. More particularly, graph 70 illustrates two shaded areas 70.sub.1 and 70.sub.2, where if the channel measurement a .alpha..sub.2,n.sup.H.alpha..sub.1,n from block 50 falls within area 70.sub.1, then block 52 computes the value of .phi..sub.2(2i) to be 0 degrees and this value is encoded by encode block 54 to produce a binary counterpart of .omega..sub.2(2i)=0 which is fed back to base station 12; conversely, if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 70.sub.2, then block 52 computes the value of .phi..sub.2(2i) to be .pi. degrees and this value is encoded by block 54 to produce a binary counterpart of .omega..sub.2(2i)=1 which is fed back to base station 12. Further, Table 3 as well as the location of points on graph 70 illustrate the phase rotation that is implemented by base station 12 in response to the possible values of {tilde over (.omega.)}.sub.2(2i). Specifically, if base station 12 receives a value of {tilde over (.omega.)}.sub.2(2i)=0, then feedback decode and process block of base station 12 treats the channel measurement phase change for slot 2i as 0 degrees; however, if base station 12 receives a value of {tilde over (.omega.)}.sub.2(2i)=1, then base station 12 treats the channel measurement phase change for slot 2i as equal to .pi. degrees.

Table 3 and FIG. 5 also illustrate that for a second slot, 2i+1, in the group i of two slots, the value of .phi..sub.2(2i+1), as determined by beamformer coefficient computation block 52 in user station 14, is based on a 90 degree rotation as shown in graph 72 and as represented by its axis 72.sub.ax which is rotated 90 degrees relative to the vertical imaginary axis. Graph 72 illustrates two shaded areas 72.sub.1 and 72.sub.2, where if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n from block 50 falls within area 72.sub.1, then block 52 computes the value of .phi..sub.2(2i+1) to be .pi./2 degrees and this value is encoded by encode block 54 into a binary counterpart .omega..sub.2(2i+1) equal to 0 which is fed back to base station 12; conversely, if the channel measurement .alpha..sub.2,n.sup.H.alpha..sub.1,n falls within area 72.sub.2, then block 52 computes the value of .phi..sub.2(2i+1) to be -.pi./2 degrees and this value is encoded by block 54 into a binary counterpart .omega..sub.2(2i+1) equal to 1 which is fed back to base station 12. Further, Table 3 as well as the location of points on graph 72 illustrate the phase rotation that is implemented by base station 12 in response to the values of {tilde over (.omega.)}.sub.2(2i+1). Specifically, if base station 12 receives a value of {tilde over (.omega.)}.sub.2 (2i+1) equal to 0, then feedback decode and process block of base station 12 treats the channel measurement phase change for slot 2i+1 as .pi./2 degrees; however, if base station 12 receives a value of {tilde over (.omega.)}.sub.2(2i+1) equal to 1, then base station