Iterative detection and decoding for a MIMO-OFDM system Number:7,154,936 from the United States Patent and Trademark Office (PTO) owispatent

Title: Iterative detection and decoding for a MIMO-OFDM system

Abstract: Techniques to iteratively detect and decode data transmitted in a wireless (e.g., MIMO-OFDM) communication system. The iterative detection and decoding is performed by iteratively passing soft (multi-bit) "a priori" information between a detector and a decoder. The detector receives modulation symbols, performs a detection function that is complementary to the symbol mapping performed at the transmitter, and provides soft-decision symbols for transmitted coded bits. "Extrinsic information" in the soft-decision symbols is then decoded by the decoder to provide its extrinsic information, which comprises the a priori information used by the detector in the detection process. The detection and decoding may be iterated a number of times. The soft-decision symbols and the a priori information may be represented using log-likelihood ratios (LLRs). Techniques are provided to reduce the computational complexity associated with deriving the LLRs, including interference nulling to isolate each transmitted signal and "dual-maxima" approximation.

Patent Number: 7,154,936 Issued on 12/26/2006 to Bjerke,   et al.

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Inventors:	Bjerke; Bjorn A. (Boston, MA), Ketchum; John W. (Harvard, MA), Walton; Jay R. (Westford, MA)
Assignee:	Qualcomm, Incorporated (San Diego, CA)
Appl. No.:	10/005,104
Filed:	December 3, 2001

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Current U.S. Class:	375/148 ; 375/149; 375/150; 714/784
Current International Class:	H04B 1/69 (20060101); H04B 1/713 (20060101)
Field of Search:	375/340,341,259,260,261,262,342,264,330,136,148,149,150 370/337,347,349,428 714/750,746,786,755,758

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References Cited [Referenced By]

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U.S. Patent Documents

	6222835		April 2001		Franz et al.
	6307882		October 2001		Marzetta
	6442214		August 2002		Boleskei et al.
	6499128		December 2002		Gerlach et al.
	6807146		October 2004		McFarland
	6856656		February 2005		Eidson et al.
	2002/0034261		March 2002		Eidson et al.
	2002/0165626		November 2002		Hammons et al.
	2003/0076890		April 2003		Hochwald et al.
	2003/0086504		May 2003		Magee et al.
	2003/0112901		June 2003		Gupta
	2004/0100897		May 2004		Shattil
	2004/0174939		September 2004		Wang
	2004/0240590		December 2004		Cameron et al.

Foreign Patent Documents

	WO 98/51111		Nov., 1998		WO
	WO 00/72496		Nov., 2000		WO
	WO 01/19013		Mar., 2001		WO
	WO 01/67702		Sep., 2001		WO

Other References

Alamouti, Siavash, "A simple Transmit Diversity Technique for Wireless Communication", IEEE, Journal on select area in communication, vol. 16, No. 8, Oct. 1998. cited by examiner . Nguyen, Anh V. "Iterative Demodulation and Decoding of Differential Space-time Block Codes", IEEE, VTS-2000, 52.sup.nd, vol. 5, pp. 2394-2400, Sep. 24-28, 2000. cited by examiner . Stefanov, A; Duman, T.M; "Turbo-coded modulation for systems with transmit and receive antenna diversity over block fading channels: system model, decoding approaches, and practical considerations", IEEE Journal on vol. 19, Issue 5, May 2001 pp. 958-968. cited by examiner . Patrick Robertson et al., "A Comparison of Optimal and Sub-Optimal MAP Decoding Algorithms Operating in the Log Domain," IEEE 1995 (pp. 1009-1013). cited by other . Andrew J. Viterbi "An Intuitive Justification and a Simplified Implementation of the MAP Decoder for Convolutional Codes," IEEE Journal on Selected Areas in Communications, vol. 16, No. 2, Feb. 1998 (pp. 260-264). cited by other . John A. C. Bingham "Multicarrier Modulation for Data Transmission: An Idea Whose Time Has Come," IEEE Communications Magazine, May 1990 (pp. 5-14). cited by other . Nguyen A. V. et al., "Iterative Demodulation and Decoding of Differential Space-Time Block Codes," IEEE Vehicular Technology Conference, VTC 2000-Fall, Conf. 52, vol. 5 of 6, Sep. 24, 2000, pp. 2394-2400. cited by other . Tonello A. M., "Iterative MAP Detection of Coded M-DPSK Signals in Fading Channels with Application to IS-136 TDMA," IEEE Vehicular Technology Conference, VTC 1999-Fall, Conf. 50. Sep. 19, 1999, pp. 1615-1619. cited by other . Petzold M. C. et al., "Performance of Interleaved Serial Concatenated Convolutional Codes with DSS/DPSK Multicarrier Modulation in a Doppler Spread Channel," IEEE Vehicular Technology Conference, May 16, 1999, pp. 77-81. cited by other . Keller T. et al., "A Turbo-Coded Burst-By-Burst Adaptive Wide-Band Speech Transceiver," IEEE Journal on Selected Areas in Communications, IEEE Inc., vol. 18, No. 11, Nov. 2000, pp. 2363-2372. cited by other . Lau V. K. N., "Variable-Rate Adaptive Channel Coding for CDMA-Reverse Link," Bell Labs Technology, Bell Laboratories, vol. 5, No. 4, Oct. 1, 2000, pp. 138-156. cited by other . Choi et al "Concatenated Space-Time Block Coded And Turbo Coded Symbol-By-Symbol Adaptive OFDM and Multi-Carrier CDMA Systems" IEEE (Jun. 2001). cited by other . Keller et al "Adaptive Modulation Techniques for Duplex OFDM Transmission" IEEE, vol. 49, No. 5 (Sep. 2000). cited by other . Golden et al "Detection algorithm and initial laboratory results using V-BLAST space-time communication architecture" Electronics Letters vol. 35, No. 1 (Jan. 7, 1999). cited by other . Ariyavisitakul, "Turbo Space-Time Processing To Improve Wireless Channel Capacity" IEEE, vol. 48, No. 8 (Aug. 2000). cited by other . Stefanov et al "Turbo-Coded Modulation for Systems With Transmit And Receive Antenna Diversity Over Block Fading Channels: System Model, Decoding Approaches, and Practical Considerations" IEEE, vol. 19, No. 5 (May 2001). cited by other . Benedetto et al "Serial Concatenation of Interleaved Codes: Performance Analysis, Design and Iterative Decoding" IEEE, vol. 44, No. 3 (May 1998). cited by other . Ten Brink, Stephan "Iterative Demapping and Decoding For Multilevel Modulation" IEEE (1998). cited by other . Tarokh et al "Combined Array Processing And Space-Time Coding" IEEE vol. 45, No. 4 (May 1999). cited by other . Bahl et al "Optimal Decoding of Linear Codes for Minimizing Symbol Error Rate" IEEE (Mar. 1974). cited by other.

Primary Examiner: Ghayour; Mohammed
Assistant Examiner: Ghulamali; Qutub
Attorney, Agent or Firm: Wadsworth; Philip R. Minhas; Sandip (Micky) S. Milikovsky; Dmitry R.

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Claims

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What is claimed is:

1. A method for recovering data transmitted in a wireless communication system, wherein the wireless communication system comprises a plurality of receive antennas, comprising: receiving at least one data stream, wherein each data stream comprises a plurality of modulation symbols for a plurality of transmitted coded bits; wherein the plurality of modulation symbols for each receive antenna is a respective modulation or coding scheme; determining for each data stream a plurality of soft decision symbols based on the received plurality of modulation symbols and a first a priori information for the transmitted coded bits; determining a second a priori information based on the plurality of soft decision symbols and the first a priori information for the transmitted coded bits; determining the first a priori information for the transmitted coded bits based in part on the second a priori information; repeating the determining the plurality of soft decision symbols and the determining the first a priori information and a plurality of times; determining decoded bits for the transmitted coded bits based on the second a priori information.

2. The method of claim 1, further comprising: deriving the first a priori information for the transmitted coded bits based on a first extrinsic information, deriving the first extrinsic information based on the second a priori information and a a posteriori information, deriving the second a priori information based on a second extrinsic information for the transmitted coded bits, and deriving the second extrinsic information based on the plurality of soft decision symbols and the first a priori information for the transmitted coded bits.

3. The method of claim 1, wherein the soft-decision symbols are represented as log-likelihood ratios (LLRs).

4. The method of claim 3, wherein a dual-maxima approximation is used to derive the LLRs for the coded bits.

5. The method of claim 1, wherein the soft-decision symbols comprise channel information.

6. The method of claim 1, wherein the soft-decision symbols comprise information for one or more spatial subchannels and one or more frequency subchannels used to transmit the plurality of modulation symbols.

7. The method of claim 2, further comprising: deinterleaving the the second extrinsic information, wherein the deinterleaved second extrinsic information is decoded; and interleaving the first extrinsic information, wherein the interleaved first extrinsic information is used to derive the soft decision symbols.

8. The method of claim 1, wherein the wireless communication system is a multiple-input multiple-output (MIMO) system.

9. The meted of claim 8, wherein the MIMO system implements orthogonal frequency division multiplexing (OFDM).

10. The method of claim 1, further comprising: recovering a first subset of the modulation symbols for a first transmit antenna by nulling a other subsets of the modulation symbols for a other transmit antennas.

11. The method of claim 10, wherein the recovering the first subset of the modulation symbols for the first transmit antenna includes pre-multiplying the received modulation symbols with a plurality of nulling matrices to derive the first subset of the recovered modulation symbols for a plurality of frequency subchannels of the first transmit antenna.

12. The method of claim 1, further comprising: recovering a first subset of the modulation symbols for a first transmit antenna by nulling the modulation symbols for a other transmit antennas from the received modulation symbols, and canceling interference due to the recovered modulation symbols from the received modulation symbols, thereby producing interference-cancelled modulation symbols, and recovering a other subsets of the modulation symbols from the interference-cancelled modulation symbols.

13. The method of claim 1, further comprising: deriving pre-decoding interference estimates based on the soft-decision symbols; and canceling the pre-decoding interference estimates from input modulation symbols, and wherein the input modulation symbols for a first transmit antenna are the received modulation symbols and the input modulation symbols for each subsequent transmit antenna are the interference-cancelled modulation symbols from the current transmit antenna.

14. The method of claim 1, wherein the soft-decision symbol for each coded bit comprises extrinsic information extracted from other coded bits.

15. The method of claim 1, wherein the decoding is based on a parallel concatenated convolutional decoding scheme.

16. The method of claim 1, wherein the decoding is based on a serial concatenated convolutional decoding scheme.

17. The method of claim 1, wherein the decoding is based on a convolutional decoding scheme.

18. The method of claim 1, wherein the decoding is based on a block decoding scheme.

19. The method of claim 1, wherein the decoding is based on a concatenated convolutional decoding scheme, and wherein a dual-maxima approximation is used for evaluating log-likelihood ratios (LLRs) for the decoding.

20. The method of claim 1, wherein the decoding for each transmit antenna is based on a respective decoding scheme.

21. The method of claim 1, wherein the plurality of modulation symbols are derived based on a non-Gray modulation scheme.

22. The method of claim 1, wherein the modulation symbols for each transmit antenna are derived based on a respective modulation scheme.

23. A receiver unit in a wireless communication system, wherein the wireless communication system comprises a plurality of receive antennas, comprising: a detector operative to receive at least one data stream, wherein each data stream comprises a plurality of modulatian symbols for a plurality of transmitted coded bits, derive soft-decision symbols for the coded bits based on the received modulation symbols and a first a priori information for the coded bits derive the first a priori information for the coded bits based on a a posteriori information, and derive a second a priori information for the coded bits based on the soft-decision symbols and the first a priori information; wherein the detector operative to receive the plurality of modulation and symbols from each receive antenna with a respective modulation or coding schemes; a decoder operative to decode the second a priori information to derive the a posteriori information and to determine decoded bits of the transmitted coded bits; and wherein the second a priori information is derived by the detector and decoded by the decoder a plurality of times prior to determining the decoded bits.

24. The receiver unit of claim 23, further comprising: a deinterleaver operative to deinterleave a second extrinsic information based on the soft-decision symbols and the first a priori information for the transmitted coded bits, wherein the deinterleaved second extrinsic information is decoded by the decoder; and an interleaver operative to interleave a first extrinsic information based on the second a priori information and the a posteriori information for the transmitted coded bits, wherein the interleaved first extrinsic information is used by the detector to derive the soft-decision symbols.

25. The receiver unit of claim 23, wherein the soft-decision symbols represent log-likelihood ratios (LLRs) for the coded bits.

26. The receiver unit of claim 25, wherein the detector is operative to use a dual-maxima approximation to derive the LLRs for the coded bits.

27. The receiver unit of claim 23, wherein the detector is further operative to recover the modulation symbols for each transmit antenna by nulling the modulation symbols for other transmit antennas, and to derive the soft-decision symbols for the coded bits transmitted from each transmit antenna based on the recovered modulation symbols for the transmit antenna and the first a priori information.

28. The receiver unit of claim 27, wherein the detector is further operative to pre-multiply the received modulation symbols with a plurality of nulling matrices to derive the recovered modulation symbols for the plurality of frequency subchannels of each transmit antenna.

29. The receiver unit of claim 27, wherein the detector is further operative to cancel interference due to the recovered modulation symbols for each transmit antenna, and to recover the modulation symbols for each subsequent transmit antenna, except the last transmit antenna, based on the interference-cancelled modulation symbols.

30. The receiver unit of claim 23, wherein one decoder is provided for each independently coded data stream to be decoded by the receiver.

31. The receiver unit of claim 23, wherein at least one decoder is operative to perform concatenated convolutional decoding on the second a priori information.

32. The receiver unit of claim 23, wherein at least one decoder implements a maximum a posteriori (MAP) decoding algorithm.

33. The receiver unit of claim 23, further comprising: a channel estimator operative to estimate one or more characteristics of a communication channel via which the plurality of modulation symbols are received; and a transmitter unit operative to process and transmit channel state information indicative of the estimated channel characteristics.

34. The receiver unit of claim 33, wherein the channel state information is indicative of a particular coding and modulation scheme to be used for each transmit antenna.

35. The receiver unit of claim 33, wherein the channel state information is indicative of a particular coding and modulation scheme to be used for all transmit antennas.

36. The receiver unit of claim 23, wherein the wireless communication system is a multiple-input multiple-output (MIMO) system that implements orthogonal frequency division multiplexing (OFDM).

37. A terminal comprising the receiver unit of claim 23.

38. A base station comprising the receiver unit of claim 23.

39. An access point comprising the receiver unit of claim 23.

40. A receiver apparatus in a wireless communication system, wherein the wireless communication system comprises a plurality of receive antennas, comprising: means for receiving at least one data stream, wherein each data stream comprises a plurality of modulation symbols for a plurality of coded bits transmitted via a plurality of frequency subchannels of the plurality of transmit antennas, wherein the plurality of modulation symbols for each receive antenna is a respective modulation or coding scheme; means for deriving soft-decision symbols for the coded bits based on the received modulation symbols and a first a priori information for the coded bits; means for deriving a second a priori information for the coded bits based on the soft-decision symbols and the first a priori information; first means for decoding the second a priori information to derive the first a priori information, wherein the second a priori information is derived and decoded a plurality of times; and first means for determining decoded bits of the transmitted coded bits based in part on the first a priori information.

41. The receiver apparatus of claim 40, further comprising: means for recovering the modulation symbols for each transmit antenna by nulling the modulation symbols for other transmit antennas, and wherein the soft-decision symbols for the coded bits transmitted from each transmit antenna are derived based on the recovered modulation symbols for the transmit antenna and the first a priori information for the transmit antenna.

42. The receiver apparatus of claim 40, further comprising: means for deinterleaving a second extrinsic information based on the soft-decision symbols and the first a priori information for the transmitted coded bits, wherein the deinterleaved second extrinsic information is decoded; and means for interleaving a first extrinsic information based on the second a priori information and a a posteriori information for the transmitted coded bits, wherein the interleaved first extrinsic information is used to derive the soft-decision symbols.

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Description

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BACKGROUND

1. Field

The present invention relates generally to data communication, and more specifically to techniques for performing iterative detection and decoding for a MIMO-OFDM communication system.

2. Background

A multiple-input multiple-output (MIMO) communication system employs multiple (N.sub.T) transmit antennas and multiple (N.sub.R) receive antennas for data transmission. A MIMO channel formed by the N.sub.T transmit and N.sub.R receive antennas may be decomposed into N.sub.S independent channels, with N.sub.S.ltoreq.min {N.sub.T, N.sub.R}. Each of the N.sub.S independent channels is also referred to as a spatial subchannel of the MIMO channel and corresponds to a dimension. The MIMO system can provide improved performance (e.g., increased transmission capacity) over that of a single-input single-output (SISO) communication system if the additional dimensionalities created by the multiple transmit and receive antennas are utilized.

A wideband MIMO system typically experiences frequency selective fading, i.e., different amounts of attenuation across the system bandwidth. This frequency selective fading causes inter-symbol interference (ISI), which is a phenomenon whereby each symbol in a received signal acts as distortion to subsequent symbols in the received signal. This distortion degrades performance by impacting the ability to correctly detect the received symbols. As such, ISI is a non-negligible noise component that may have a large impact on the overall signal-to-noise-and-interference ratio (SNR) for systems designed to operate at high SNR levels, such as MIMO systems. In such systems, equalization may be used at the receivers to combat ISI. However, the computational complexity required to perform equalization is typically significant or prohibitive for most applications.

Orthogonal frequency division multiplexing (OFDM) may be used to combat ISI, and achieves this without the use of computationally intensive equalization. An OFDM system effectively partitions the system bandwidth into a number of (N.sub.F) frequency subchannels, which may be referred to as sub-bands or frequency bins. Each frequency subchannel is associated with a respective subcarrier upon which data may be modulated. The frequency subchannels of the OFDM system may experience frequency selective fading (i.e., different amounts of attenuation for different frequency subchannels), depending on the characteristics (e.g., multipath profile) of the propagation path between the transmit and receive antennas. With OFDM, the ISI due to the frequency selective fading may be combated by repeating a portion of each OFDM symbol (i.e., appending a cyclic prefix to each OFDM symbol), as is known in the art.

A MIMO system may thus advantageously employ OFDM to combat ISI. The frequency subchannels of the MIMO-OFDM system may experience different channel conditions (e.g., different fading and multipath effects) and may achieve different SNRs. Moreover, the channel conditions may vary over time. Consequently, the supported data rates may vary from frequency subchannel to frequency subchannel and from spatial subchannel to spatial subchannel, and may further vary with time. To achieve high performance, it is necessary to properly code and modulate the data at the transmitter (e.g., based on the determined channel conditions) and to properly detect and decode the received signals at the receiver.

There is therefore a need in the art for techniques to detect and decode signals that may have been (flexibly) coded and modulated based on one or more coding and modulation schemes, e.g., as determined by the channel conditions.

SUMMARY

Aspects of the invention provide techniques to iteratively detect and decode data transmitted in a wireless (e.g., MIMO-OFDM) communication system. The iterative detection and decoding exploits the error correction capabilities of the channel code to provide improved performance. This is achieved by iteratively passing soft (multi-bit) "a priori" information between a soft-input soft-output detector and a soft-input soft-output decoder.

The detector receives modulation symbols previously generated at a transmitter system based on one or more coding and modulation schemes, performs a detection function that is complementary to the symbol mapping performed at the transmitter system, and provides soft-decision symbols for transmitted coded bits. Extrinsic information in the soft-decision symbols (which comprises the a priori information for the decoder, as described below) is then decoded by the decoder based on one or more decoding schemes complementary to the one or more coding schemes used at the transmitter system. The decoder further provides its extrinsic information (which comprises the a priori information for the detector) that is then used by the detector in the detection process.

The detection and decoding may be iterated a number of times. During the iterative detection and decoding process, the reliability of the bit decisions is improved with each iteration. The iterative detection and decoding process described herein may be used to combat frequency selective fading as well as flat fading. Moreover, the iterative detection and decoding process may be flexibly used with various types of coding schemes (e.g., serial and parallel concatenated convolutional codes) and with various modulation schemes (e.g., M-PSK and M-QAM).

The a priori information passed between the detector and decoder and the soft-decision symbols may be represented using log-likelihood ratios (LLRs). Techniques are provided herein to reduce the computational complexity associated with deriving the LLRs. Such techniques include the use of interference nulling to isolate each transmitted signal by removing the other interferers and the use of a "dual-maxima" or some other approximation to compute the LLRs, which are described below.

Various aspects and embodiments of the invention are described in further detail below. The invention further provides methods, receiver units, transmitter units, receiver systems, transmitter systems, systems, and other apparatuses and elements that implement various aspects, embodiments, and features of the invention, as described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, nature, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein:

FIG. 1 is a block diagram of a transmitter system and a receiver system in a MIMO-OFDM system;

FIGS. 2A and 2B are block diagrams of two transmitter units that code and modulate data with (1) a single coding and modulation scheme and (2) separate coding and modulation schemes on a per-antenna basis, respectively;

FIGS. 3A and 3B are block diagrams of serial and parallel concatenated convolutional encoders, respectively;

FIG. 3C is a block diagram of a recursive convolutional encoder;

FIGS. 4A and 4B are block diagrams of two receiver units that detect and decode data previously processed with (1) a single coding and modulation scheme and (2) separate coding and modulation schemes on a per-antenna basis, respectively;

FIG. 4C is a block diagram of a receiver unit that performs successive nulling and interference cancellation to recover one transmitted signal at a time;

FIGS. 5A and 5B are block diagrams of two Turbo decoders capable of performing iterative decoding for serial and parallel concatenated convolutional codes, respectively; and

FIG. 6 is a block diagram of an interference canceller that may be used for the receiver unit in FIG. 4C.

DETAILED DESCRIPTION

The iterative detection and decoding techniques described herein may be used for various wireless communication systems. For clarity, various aspects and embodiments of the invention are described specifically for multiple-input multiple output communication system that implements orthogonal frequency division multiplexing (i.e., a MIMO-OFDM system).

As noted above, a MIMO system employs N.sub.T transmit antennas and N.sub.R receive antennas for data transmission, where N.sub.R.gtoreq.N.sub.T. A MIMO channel formed by the N.sub.T transmit antennas and N.sub.R receive antennas may be decomposed into N.sub.S spatial subchannels, where N.sub.S.ltoreq.min {N.sub.T, N.sub.R}. An OFDM system effectively partitions the system bandwidth into N.sub.F frequency subchannels. Each frequency subchannel may be defined to be sufficiently narrow so that its frequency response is considered flat or frequency non-selective. A MIMO-OFDM system may thus transmit data via a number of (N.sub.C) "transmission channels" (where N.sub.C=N.sub.SN.sub.F), with each such transmission channel corresponding to a frequency subchannel of a spatial subchannel.

FIG. 1 is a block diagram of an embodiment of a transmitter system 110 and a receiver system 150 in a MIMO-OFDM system 100. Transmitter system 110 and receiver system 150 are capable of implementing various aspects and embodiments of the invention, as described below.

At transmitter system 110, traffic data is provided at a particular data rate from a data source 112 to a transmit (TX) data processor 114, which codes and interleaves the traffic data based on one or more coding schemes to provide coded data. The coding may be performed based on a single coding scheme for all transmit antennas, one coding scheme for each transmit antenna or each subset of transmit antennas, or one coding scheme for each transmission channel or each group of transmission channels. The data rate and the coding may be determined by a data rate control and a coding control, respectively, provided by a controller 130.

The coded data is then provided to a modulator 116, which may also receive pilot data (e.g., data of a known pattern and processed in a known manner). The pilot data may be multiplexed with the coded traffic data (e.g., using time division multiplexing (TDM) or code division multiplexing (CDM)) in all or a subset of the frequency subchannels and in all or a subset of the spatial subchannels used to transmit the traffic data. The pilot may be used by the receiver system to perform a number of functions such as acquisition, frequency and timing synchronization, channel estimation, coherent data demodulation, and so on.

In a specific embodiment, the processing by modulator 116 includes (1) modulating the received data with one or more modulation schemes (e.g., M-PSK, M-QAM, and so on) to provide modulation symbols, (2) transforming the modulation symbols to form OFDM symbols, and (3) appending a cyclic prefix to each OFDM symbol to form a corresponding transmission symbol. Similarly, the modulation may be performed based on a single modulation scheme for all transmit antennas, one modulation scheme for each transmit antenna or each subset of transmit antennas, or one modulation scheme for each transmission channel or each group of transmission channels. The modulation is performed based on a modulation control provided by controller 130. The modulated data (i.e., the transmission symbols) is then provided to transmitters (TMTR) 122a through 122t associated with the N.sub.T transmit antennas to be used for data transmission.

Each transmitter 122 converts the received modulated data into one or more analog signals and further conditions (e.g., amplifies, filters, and quadrature modulates) the analog signals to generate a modulated signal suitable for transmission over the communication channel. The modulated signals from transmitters 122a through 122t are then transmitted via antennas 124a through 124t, respectively, to the receiver system.

At receiver system 150, the transmitted modulated signals are received by antennas 152a through 152r, and the received signal from each antenna is provided to a respective receiver (RCVR) 154. Each receiver 154 conditions (e.g., filters, amplifies, and downconverts) a respective received signal and digitizes the conditioned signal to provide a respective stream of data samples, which represent the transmission symbols received via the associated antenna. A demodulator (Demod) 156 receives and demodulates the N.sub.R data sample streams from receivers 154a through 154r to provide N.sub.R corresponding streams of received modulation symbols. For each data sample stream, demodulator 156 removes the cyclic prefix included in each transmission symbol and then transforms each received OFDM symbol to provide a corresponding stream of received modulation symbols.

A detector/decoder 158 initially performs the detection function that is complementary to the symbol mapping and provides soft-decision (multi-bit) symbols for the coded bits transmitted from the transmitter system. The soft-decision symbols are then decoded based on one or more decoding schemes complementary to the one or more coding schemes used at the transmitter system. In an aspect, the detection and decoding may be performed iteratively a number of times, as described in further detail below. The decoded data is then provided to a data sink 160.

Controllers 130 and 170 direct the operation at the transmitter and receiver systems, respectively. Memories 132 and 172 provide storage for program codes and data used by controllers 130 and 170, respectively.

Transmitter System

FIG. 2A is a block diagram of a transmitter unit 200a, which is an embodiment of the transmitter portion of transmitter system 110 in FIG. 1. In this embodiment, a single coding scheme is used for all N.sub.T transmit antennas and a single modulation scheme is used for all N.sub.F frequency subchannels of all transmit antennas. Transmitter unit 200a includes (1) a TX data processor 114a that receives and codes traffic data in accordance with a specific coding scheme to provide coded data and (2) a modulator 116a that modulates the coded data in accordance with a specific modulation scheme to provide modulated data. TX data processor 114a and modulator 116a are thus one embodiment of TX data processor 114 and modulator 116, respectively, in FIG. 1.

In the specific embodiment shown in FIG. 2A, TX data processor 114a includes an encoder 212, a channel interleaver 214, and a demultiplexer (Demux) 216. Encoder 212 receives and codes the traffic data (i.e., the information bits) in accordance with the selected coding scheme to provide coded bits. The coding increases the reliability of the data transmission. The selected coding scheme may include any combination of cyclic redundancy check (CRC) coding, convolutional coding, Turbo coding, block coding, and so on. Several designs for encoder 212 are described below.

Channel interleaver 214 then interleaves the coded bits based on a particular interleaving scheme and provides interleaved coded bits. The interleaving provides time diversity for the coded bits, permits the data to be transmitted based on an average signal-to-noise-and-interference ratio (SNR) for the frequency and/or spatial subchannels used for the data transmission, combats fading, and further removes correlation between coded bits used to form each modulation symbol. The interleaving may further provide frequency diversity if the coded bits are transmitted over multiple frequency subchannels. The coding and channel interleaving are described in further detail below.

Demultiplexer 216 then demultiplexes the interleaved and coded data into N.sub.T coded data streams for the N.sub.T transmit antennas to be used for the data transmission. The N.sub.T coded data streams are then provided to modulator 116a.

In the specific embodiment shown in FIG. 2A, modulator 116a includes N.sub.T OFDM modulators, with each OFDM modulator assigned to process a respective coded data stream for one transmit antenna. Each OFDM modulator includes a symbol mapping element 222, an inverse fast Fourier transformer (IFFT) 224, and a cyclic prefix generator 226. In this embodiment, all N.sub.T symbol mapping elements 222a through 222t implement the same modulation scheme.

Within each OFDM modulator, symbol mapping element 222 maps the received coded bits to modulation symbols for the (up to) N.sub.F frequency subchannels to be used for data transmission on the transmit antenna associated with the OFDM modulator. The particular modulation scheme to be implemented by symbol mapping element 222 is determined by the modulation control provided by controller 130. For OFDM, the modulation may be achieved by grouping sets of q coded bits to form non-binary symbols and mapping each non-binary symbol to a specific point in a signal constellation corresponding to the selected modulation scheme (e.g., QPSK, M-PSK, M-QAM, or some other scheme). Each mapped signal point corresponds to an M-ary modulation symbol, where M=2.sup.q. Symbol mapping element 222 then provides a vector of (up to) N.sub.F modulation symbols for each transmission symbol period, with the number of modulation symbols in each vector corresponding to the number of frequency subchannels to be used for data transmission for that transmission symbol period.

If conventional non-iterative symbol de-mapping and decoding are performed at the receiver system, then Gray mapping may be preferably used for the symbol mapping since it may provide better performance in terms of bit error rate (BER). With Gray mapping, the neighboring points in the signal constellation (in both the horizontal and vertical directions) differ by only one out of the q bit positions. Gray mapping reduces the number of bit errors for more likely error events, which correspond to a received modulation symbol being mapped to a location near the correct location, in which case only one coded bit would be received in error.

However, if iterative detection and decoding are performed as described below, it can be shown that non-Gray mapping outperforms Gray mapping. This is true due to the fact that independence between the coded bits enhances independence between the detection and decoding processes, which then provides improved performance when iterative detection and decoding are performed. Thus, each symbol mapping element 222 may be designed to implement a non-Gray mapped constellation. In certain instances, improved performance may be achieved if the constellation is defined such that neighboring points differ by as many bit positions as possible (i.e., the opposite goal as for Gray mapping, or "anti-Gray" mapping).

IFFT 224 then converts each modulation symbol vector into its time-domain representation (which is referred to as an OFDM symbol) using the inverse fast Fourier transform. IFFT 224 may be designed to perform the inverse transform on any number of frequency subchannels (e.g., 8, 16, 32, . . . , N.sub.F, . . . ). In an embodiment, for each OFDM symbol, cyclic prefix generator 226 repeats a portion of the OFDM symbol to form a corresponding transmission symbol. The cyclic prefix ensures that the transmission symbol retains its orthogonal properties in the presence of multipath delay spread, thereby improving performance against deleterious path effects such as channel dispersion caused by frequency selective fading. The transmission symbols from cyclic prefix generator 226 are then provided to an associated transmitter 122 and processed to generate a modulated signal, which is then transmitted from the associated antenna 124.

FIG. 2B is a block diagram of a transmitter unit 200b, which is another embodiment of the transmitter portion of transmitter system 110 in FIG. 1. In this embodiment, a particular coding scheme is used for each of the N.sub.T transmit antennas and a particular modulation scheme is used for all N.sub.F frequency subchannels of each transmit antenna (i.e., separate coding and modulation on a per-antenna basis). The specific coding and modulation schemes to be used for each transmit antenna may be selected based on the expected channel conditions (e.g., by the receiver system and sent back to the transmitter system).

Transmitter unit 200b includes (1) a TX data processor 114b that receives and codes traffic data in accordance with separate coding schemes to provide coded data and (2) a modulator 116b that modulates the coded data in accordance with separate modulation schemes to provide modulated data. TX data processor 114b and modulator 116b are another embodiment of TX data processor 114 and modulator 116, respectively, in FIG. 1.

In the specific embodiment shown in FIG. 2B, TX data processor 114b includes a demultiplexer 210, N.sub.T encoders 212a through 212t, and N.sub.T channel interleavers 214a through 214t (i.e., one set of encoder and channel interleaver for each transmit antenna). Demultiplexer 210 demultiplexes the traffic data (i.e., the information bits) into N.sub.T data streams for the N.sub.T transmit antennas to be used for the data transmission. Each data stream is then provided to a respective encoder 212.

Each encoder 212 receives and codes a respective data stream based on the specific coding scheme selected for the corresponding transmit antenna to provide coded bits. The coded bits from each encoder 212 are then provided to a respective channel interleaver 214, which interleaves the coded bits based on a particular interleaving scheme to provide diversity. Channel interleavers 214a through 214t then provide to modulator 116b N.sub.T interleaved and coded data streams for the N.sub.T transmit antennas.

In the specific embodiment shown in FIG. 2B, modulator 116b includes N.sub.T OFDM modulators, with each OFDM modulator including symbol mapping element 222, IFFT 224, and cyclic prefix generator 226. In this embodiment, the N.sub.T symbol mapping elements 222a through 222t may implement different modulation schemes. Within each OFDM modulator, symbol mapping element 222 maps groups of q.sub.n coded bits to form M.sub.n-ary modulation symbols, where M.sub.n corresponds to the specific modulation scheme selected for the n-th transmit antenna (as determined by the modulation control provided by controller 130) and M.sub.n=2.sup.qn. The subsequent processing by IFFT 224 and cyclic prefix generator 226 is as described above.

Other designs for the transmitter unit may also be implemented and are within the scope of the invention. For example, the coding and modulation may be separately performed for each subset of transmit antennas, each transmission channel, or each group of transmission channels. The implementation of encoders 212, channel interleavers 214, symbol mapping elements 222, IFFTs 224, and cyclic prefix generators 226 is known in the art and not described in detail herein.

The coding and modulation for MIMO systems with and without OFDM are described in further detail in U.S. patent application Ser. Nos. 09/826,481 and 09/956,449, both entitled "Method and Apparatus for Utilizing Channel State Information in a Wireless Communication System," respectively filed Mar. 23, 2001 and Sep. 18, 2001; U.S. patent application Ser. No. 09/854,235, entitled "Method and Apparatus for Processing Data in a Multiple-Input Multiple-Output (MIMO) Communication System Utilizing Channel State Information," filed May 11, 2001; U.S. patent application Ser. No. 09/776,075, entitled "Coding Scheme for a Wireless Communication System," filed Feb. 1, 2001; and U.S. patent application Ser. No. 11/332,059, entitled "Multiple-Access Multiple-Input Multiple-Output (MIMO) Communication System," filed Nov. 6, 2001. These applications are all assigned to the assignee of the present application and incorporated herein by reference. Still other coding and modulation schemes may also be used, and this is within the scope of the invention.

An example OFDM system is described in U.S. patent application Ser. No. 09/532,492, entitled "High Efficiency, High Performance Communication System Employing Multi-Carrier Modulation," filed Mar. 30, 2000, assigned to the assignee of the present invention and incorporated herein by reference. OFDM is also described by John A. C. Bingham in a paper entitled "Multicarrier Modulation for Data Transmission: An Idea Whose Time Has Come," IEEE Communications Magazine, May 1990, which is incorporated herein by reference.

Encoding

Various types of encoder may be used to code data prior to transmission. For example, the encoder may implement any one of the following (1) a serial concatenated convolutional code (SCCC), (2) a parallel concatenated convolutional code (PCCC), (3) a simple convolutional code, (4) a concatenated code comprised of a block code and a convolutional code, and so on. Concatenated convolutional codes are also referred to as Turbo codes.

FIG. 3A is a block diagram of an embodiment of a serial concatenated convolutional encoder 212x, which may be used for each of encoders 212 in FIGS. 2A and 2B. Encoder 212x includes an outer convolutional encoder 312a, a code interleaver 314, and an inner convolutional encoder 312b, all coupled in series. Outer convolutional encoder 312a codes the information bits with a particular outer code of code rate R.sub.o. The coded output from encoder 312a is provided to code interleaver 314, which interleaves each packet of N.sub.P coded bits in accordance with a particular (e.g., pseudo-random) interleaving scheme.

Code interleaver 314 may implement any one of a number of interleaving schemes, such as the ones used for cdma2000 and W-CDMA. In one specific interleaving scheme, the N.sub.P coded bits in a packet are written, by row, into a 2.sup.5-row by 2.sup.n-column array, where n is the smallest integer such that N.sub.P.ltoreq.2.sup.5+n. The rows are then shuffled in accordance with a bit-reversal rule. For example, row 1 ("00001") is swapped with row 16 ("10000"), row 3 ("00011") is swapped with row 24 ("11000"), and so on. The bits within each row are then permutated (i.e., rearranged) according to a row-specific linear congruential sequence (LCS). The LCS for row k may be defined as x.sub.k(i+1)={x.sub.k(i)+c.sub.k} mod 2.sup.n, where i=0, 1, . . . 2.sup.n-1, x.sub.k(0)=c.sub.k, and c.sub.k is a specific value selected for each row and is further dependent on the value for n. For the permutation in each row, the i-th bit in the row is placed in location x(i). The bits in the array are then read out by column.

The LCS code interleaving scheme is described in further detail in commonly assigned U.S. patent application Ser. No. 09/205,511, entitled "Turbo Code Interleaver Using Linear Congruential Sequences," filed Dec. 4, 1998, and in a cdma2000 document entitled "C.S0002-A-1 Physical Layer Standard for cdma2000 Spread Spectrum Systems," both of which are incorporated herein by reference. Other code interleavers may also be used and are within the scope of the invention. For example, a random interleaver or a symmetrical-random (S-random) interleaver may also be used instead of the LCS interleaver described above.

Inner convolutional encoder 312b receives and further codes the interleaved bits from code interleaver 314 with a particular inner code of code rate R.sub.i. In an embodiment, encoder 312b implements a recursive code to fully realize the benefit of the significant interleaving gain provided by code interleaver 314. The inner code does not need to be a powerful code since the key desired property is recursiveness. In fact, the inner code may simply be a rate-1 differential code. The overall code rate for serial concatenated convolutional encoder 212x is R.sub.SCCC=R.sub.oR.sub.i.

FIG. 3B is a block diagram of an embodiment of a parallel concatenated convolutional encoder 212y, which may also be used for each of encoders 212 in FIGS. 2A and 2B. Encoder 212y includes two constituent convolutional encoder 312c and 312d, a code interleaver 324, a puncturing element 326, and a parallel-to-serial (P/S) converter 328. Code interleaver 324 interleaves the information bits in accordance with a particular (i.e., pseudo-random) interleaving scheme, and may be implemented as described above for code interleaver 314.

As shown in FIG. 3B, the information bits are provided to convolutional encoder 312c and the interleaved information bits are provided to convolutional encoder 312d. Each encoder 312 codes the received bits based on a particular constituent code and provides a respective stream of parity bits. Encoders 312c and 312d may be implemented with two recursive systematic constituent codes with code rates of R.sub.1 and R.sub.2, respectively. The recursive codes maximize the benefits provided by the interleaving gain.

The parity bits b.sup.y and b.sup.z from encoders 312c and 312d, respectively, are provided to puncturing element 326, which punctures (i.e., deletes) zero or more of the parity bits to provide the desired number of output bits. Puncturing element 326 is an optional element that may be used to adjust the overall code rate, R.sub.PCCC, of the parallel concatenated convolutional encoder, which is given by 1/R.sub.PCCC=1/R.sub.1+1/R.sub.2-1.

The information bits (which are also referred to as the systematic bits), and the punctured parity bits from convolutional encoders 312c and 312d are provided to P/S converter 328 and serialized into a coded bit stream that is provided to the next processing element.

FIG. 3C is a block diagram of an embodiment of a recursive convolutional encoder 312x, which may be used for each of encoders 312a through 312d in FIGS. 3A and 3B. Encoder 312x may also be used for each of encoders 212 in FIGS. 2A and 2B.

In the embodiment shown in FIG. 3C, encoder 312x implements the following transfer function for the recursive convolutional code:

.times..times..times..times. ##EQU00001## where n(D)=1+D+D.sup.3, and d(D)=1+D.sup.2+D.sup.3. Encoder 312x may also be designed to implement other convolutional codes, and this is within the scope of the invention.

Encoder 312x includes a number of series-coupled delay elements 332, a number of modulo-2 adders 334, and a switch 336. Initially, the states of delay elements 332 are set to zeros and switch 336 is in the up position. Then, for each received bit in a packet, adder 334a performs modulo-2 addition of the received bit with the output bit from adder 334c and provides the result to delay element 332a. Adder 334b performs modulo-2 addition of the bits from adder 334a and delay elements 332a and 332c and provides the parity bit. Adder 334c performs modulo-2 addition of the bits from delay elements 332b and 332c.

After all N.sub.I information bits in the packet have been coded, switch 336 is moved to the down position and three zero ("0") bits are provided to encoder 312x. Encoder 312x then codes the three zero bits and provides three tail systematic bits and three tail parity bits.

It can be shown analytically and via computer simulations that SCCCs provide better performance than PCCCs in additive white Gaussian noise (AWGN) channels at medium to high SNR levels, which is typically the desired operating region for MIMO systems. While the BER for PCCCs asymptotically reaches an error floor, this floor is absent or much lower for SCCCs. PCCCs outperform SCCCs in the high BER region, and may be more suitably used when the system loads approach the capacity limits of the channel at low SNRs. Both PCCCs and SCCCs may be implemented using relatively simple constituent codes (e.g., having constraint lengths of 3 to 16), such as the one shown in FIG. 3C.

Channel Interleaving

Referring back to FIGS. 2A and 2B, the coded bits from each encoder 212 are interleaved by a respective channel interleaver 214 to provide temporal, frequency, and/or spatial diversity against deleterious path effects (e.g., fading and multipath). Moreover, since the coded bits are subsequently grouped together to form non-binary symbols that are then mapped to M-ary modulation symbols, the interleaving may be used to ensure that the coded bits that form each modulation symbol are not located close to each other temporally (i.e., the channel interleaving distributes the coded bits that are temporally close together in a pseudo-random manner among modulation symbols that may be transmitted over different frequency subchannels, spatial subchannels, and/or transmission symbol periods). The combination of encoding, channel interleaving and symbol mapping (especially anti-Gray mapping) may be viewed as a serial concatenated code, where the symbol mapper takes on the role of the inner code. The channel interleaver provides interleaving gain in much the same way as in an SCCC, as described earlier. This potential for performance gain is unlocked by the iterative receiver structure described below. The channel interleaving can provide improved performance for various coding and modulation schemes, such as a single common coding and modulation scheme for all transmit antennas or separate coding and modulation scheme per antenna.

Various interleaving schemes may be used for the channel interleaver. In one interleaving scheme, the coded bits for each packet are written (linearly) to rows of an array. The bits in each row may then be permutated (i.e., rearranged) based on (1) a bit-reversal rule, (2) a linear congruential sequence (such as the one described above for the code interleaver), (3) a randomly generated pattern, or (4) a permutation pattern generated in some other manner. The rows are also permutated in accordance with a particular row permutation pattern. The permutated coded bits are then retrieved from each column of the array and provided to the next processing element. Other channel interleaving schemes may also be used and this is within the scope of the invention.

In an embodiment, the channel interleaving is performed separately for each independently coded data stream. For the PCCCs, the information bits and the tail and parity bits for each packet may also be channel interleaved separately. For example, the information bits b.sup.x, the tail and parity bits b.sup.y from the first constituent encoder 312c, and the tail and parity bits b.sup.z from the second constituent encoder 312d may be interleaved by three separate channel interleavers, which may employ the same or different interleaving schemes. This separate channel interleaving allows for flexible puncturing of the individual parity bits.

The interleaving interval may be selected to provide the desired temporal, frequency, and/or spatial diversity, or any combination thereof. For example, the coded bits for a particular time period (e.g., 10 msec, 20 msec, and so on) and for a particular combination of transmission channels may be interleaved. The channel interleaving may be performed for each transmit antenna, or across each group of transmit antennas or across all transmit antennas to provide spatial diversity. The channel interleaving may also be performed for each frequency subchannel, or across each group of frequency subchannels or across all frequency subchannels to provide frequency diversity. The channel interleaving may also be performed across each group of one or more frequency subchannels of each group of one or more transmit antennas such that the coded bits from one data stream may be distributed over one or more frequency subchannels of one or more transmit antennas to provide a combination of temporal, frequency, and spatial diversity. The channel interleaving may also be performed across all frequency subchannels of all transmit antennas.

Receiver System

FIG. 4A is a block diagram of an embodiment of a receiver unit 400a, which is an embodiment of the receiver portion of receiver system 150 in FIG. 1. In this embodiment, a single demodulation scheme is used for all N.sub.F frequency subchannels of all N.sub.T transmit antennas and a single decoding scheme is used for all transmit antennas. Receiver unit 400a may thus be used to receive a data transmission from transmitter unit 200a in FIG. 2A.

The signals transmitted from the N.sub.T transmit antennas are initially received by each of N.sub.R antennas 152a through 152r and routed to a respective receiver 154 (which is also referred to as a front-end unit). Each receiver 154 conditions (e.g., filters, amplifies, and downconverts) a respective received signal and further digitizes the conditioned signal to provide data samples. Each receiver 154 may further demodulate the data samples with a recovered pilot to provide a stream of received transmission symbols, which is provided to a demodulator 156a.

In the specific embodiment shown in FIG. 4A, demodulator 156a includes N.sub.R OFDM demodulators, with each OFDM demodulator assigned to process a respective transmission symbol stream from one receive antenna. Each OFDM demodulator includes a cyclic prefix remover 412 and a fast Fourier transformer (FFT) 414. Cyclic prefix remover 412 removes the cyclic prefix previously appended to each OFDM symbol by the transmitter system to ensure ISI-free reception of the transmitted modulation symbols. FFT 414 then transforms each received OFDM symbol to provide a vector of N.sub.F received modulation symbols for the N.sub.F frequency subchannels used to transmit the OFDM symbol. The N.sub.R modulation symbol vectors from all N.sub.R OFDM demodulators for each transmission symbol period are provided to a detector/decoder 158a, which is one embodiment of detector/decoder 158 in FIG. 1.

In the embodiment shown in FIG. 4A, detector/decoder 158a includes a detector 420a and a decoder 430 that perform iterative detection and decoding on the modulation symbols received from all N.sub.R receive antennas to provide decoded data. The iterative detection and decoding exploits the error correction capabilities of the channel code to provide improved performance. This is achieved by iteratively passing soft "a priori" information between the soft-input soft-output (SISO) detector 420a and the soft-input soft-output decoder 430, as described in further detail below.

Detector 420a receives the modulation symbols from demodulator 156a and a priori information from decoder 430 and derives soft-decision (i.e., multi-bit) symbols for all N.sub.F frequency subchannels of all N.sub.T transmit antennas, with each such soft-decision symbol being an estimate of a coded bit transmitted by the transmitter system. As described in further detail below, the soft-decision symbols may be represented as log-likelihood ratios (LLRs), which are denoted as L(b.sub.k) in FIG. 4A.

For each transmission symbol period, detector 420a provides up to N.sub.B soft-decision symbols to N.sub.B respective summers 422, where N.sub.B=N.sub.TN.sub.Fq and q is dependent on the specific modulation scheme used for the data transmission. Each summer 422 also receives the a priori information for its coded bit b.sub.k from decoder 430 (which is referred to as the detector a priori information and denoted as L.sub.a(b.sub.k)), and subtracts this detector a priori information from the received soft-decision symbol to derive extrinsic information for the coded bit (denoted as L.sub.e(b.sub.k)). The extrinsic information for all (N.sub.TN.sub.Fq) coded bits is then (1) converted from parallel to serial by a P/S converter 424, (2) deinterleaved by a channel deinterleaver 426 in a manner complementary to the channel interleaving performed at the transmitter system, and (3) provided as a priori information from the detector to the decoder (which is referred to as the decoder a priori information and denoted as L.sub.a.sup.D(b.sub.k)).

Decoder 430 uses the decoder a priori information in the decoding process and provides the decoded data. Decoder 430 further provides "a posteriori" information (denoted as L.sup.D(b.sub.k)) to