Title: Method for adjusting a phase angle of a phase modifier of a transmitting device
Abstract: A method for adjusting a phase angle (φ) of a phase modifier (25) of a transmitting device which includes a quadrature modulator (3), a power amplifier (9), a quadrature demodulator (19) and differential amplifiers (26, 27). The power amplifier (9) is linearized via the feedback loop (16) according to the Cartesian feedback method. The phase modifier (25) supplies an oscillator signal to the quadrature demodulator (19), which signal is shifted by the phase angle (φ) to be adjusted with regard to the oscillator signal that is supplied to the quadrature modulator (3). An input signal with a constant inphase component (I) and a constant quadrature phase component (Q) is applied during each transmission burst in the instance of a closed feedback loop, and the quadrature component (VQM) and/or the inphase component (VIM) is measured at a measuring point (53, 61) located behind the output of the differential amplifiers (26, 27).
Patent Number: 7,020,216 Issued on 03/28/2006 to Lipp
| Inventors:
|
Lipp; Friedrich (Hof bei Salzburg, AT)
|
| Assignee:
|
Rohde & Schwarz GmbH & Co KG (Munich, DE)
|
| Appl. No.:
|
070795 |
| Filed:
|
June 29, 2000 |
| PCT Filed:
|
June 29, 2000
|
| PCT NO:
|
PCT/EP00/06078
|
| 371 Date:
|
March 8, 2002
|
| 102(e) Date:
|
March 8, 2002
|
| PCT PUB.NO.:
|
WO01/24467 |
| PCT PUB. Date:
|
April 5, 2001 |
Foreign Application Priority Data
| Sep 29, 1999[DE] | 199 46 669 |
| Current U.S. Class: |
375/308; 375/329; 375/279; 455/24; 455/69; 455/126; 330/260; 330/294 |
| Current Intern'l Class: |
H04L 27/20 (20060101) |
| Field of Search: |
375/308,297,219,296,279,329
330/2,129,294,260
455/126,78,69,24
|
References Cited [Referenced By]
U.S. Patent Documents
| 5381108 | Jan., 1995 | Whitmarsh et al.
| |
| 5623226 | Apr., 1997 | Whitmarsh et al.
| |
| 5793817 | Aug., 1998 | Wilson.
| |
| 5894496 | Apr., 1999 | Jones.
| |
| 6466628 | Oct., 2002 | Kim.
| |
| Foreign Patent Documents |
| 0 706 259 | Apr., 1996 | EP.
| |
| WO 98/0090/8 | Jan., 1998 | WO.
| |
| WO 99/0448/6 | Jan., 1999 | WO.
| |
| WO 99/1743/9 | Jan., 1999 | WO.
| |
| WO 00/2542/1 | May., 2000 | WO.
| |
Primary Examiner: Chin; Stephen
Assistant Examiner: Wang; Ted
Attorney, Agent or Firm: Caesar, Rivise, Bernstein, Cohen & Pokotilow, Ltd.
Claims
The invention claimed is:
1. A method for adjusting a phase angle of a phase modifier of a transmitting
device, wherein said method comprises:
providing the transmitting device comprising:
a quadrature modulator for quadrature modulation of an inphase component and
a quadrature phase component of a complex input signal;
a power amplifier, connected downstream of the quadrature modulator;
a quadrature demodulator for quadrature demodulation of an output signal of the
power amplifier into a fedback inphase component and a fedback quadrature phase component;
a first differential amplifier, connected upstream the quadrature modulator,
said first differential amplifier having a first input supplied by the inphase
component of the complex input signal and a second input supplied by the fedback
inphase component;
a second differential amplifier, connected upstream the quadrature modulator,
said second differential amplifier having a first input of the second differential
amplifier supplied by the quadrature phase component of the complex input signal
and a second input of the second differential amplifier supplied by the fedback
quadrature phase component; and
a phase modifier, which supplies to the quadrature demodulator an oscillator
signal, shifted with regard to an oscillator signal supplied to the quadrature
modulator by the phase angle to be adjusted;
applying an input signal with a predetermined constant inphase component and
a predetermined constant quadrature phase component at each transmitting interval
with a closed feedback loop containing the quadrature modulator, the power amplifier,
the quadrature demodulator, the first differential amplifier and the second differential amplifier;
measuring the quadrature phase component and optionally the inphase component
at a first measuring point behind an output of the first differential amplifier
and a second measuring point behind an output of the second differential amplifier;
determining a phase correction value based on the measured quadrature phase component
and optionally the measured inphase component; and
correcting the currently set phase angle of the phase modifier by adding or subtracting
the determined phase correction value in a transmitting interruption interval,
wherein the phase angle is not altered if the amount of the measured quadrature
phase component is smaller than a predetermined limit value.
2. The method according to claim 1, wherein the quadrature phase component of
the complex input signal being applied has a value of zero, and the measuring at
the second measuring point behind the output of the second differential amplifier
takes place at a beginning of every transmitting interval.
3. The method according to claim 1, wherein the phase correction value (Δφ)
is determined by solving the following equation:
Δφ=arc tan (
VQM/VIM)-arc tan (
Q/I)
wherein V
QM is the measured quadrature phase component, V
IM is
the measured inphase component, Q is the predetermined quadrature phase component
and I is the predetermined inphase component.
4. The method according to claim 3, wherein prior to or concurrent with activating
the transmitting device, the phase angle of the phase modifier is preliminarily
adjusted such that an output power is measured at a power detector connected downstream
of the power amplifier and the phase angle is pre-adjusted such that a minimum
of the output power results.
5. The method according to claim 1, wherein the determining of the phase correction
value comprises altering the phase angle by a step width in a first direction if
the measured quadrature phase component is positive and altering the phase angle
by a step width in an opposite direction if the measured quadrature phase component
is negative.
6. The method according to claim 5, wherein the step width depends on an amount
of the measured quadrature component.
7. The method according to claim 1, wherein prior to or concurrent with activating
the transmitting device, the phase angle of the phase modifier is preliminarily
adjusted such that an output power is measured at a power detector connected downstream
of the power amplifier and the phase angle is pre-adjusted such that a minimum
of the output power results.
8. A method for adjusting a phase angle of a phase modifier of a transmitting
device, wherein said method comprises:
providing the transmitting device comprising:
a quadrature modulator for quadrature modulation of an inphase component and
a quadrature phase component of a complex input signal;
a power amplifier, connected downstream of the quadrature modulator;
a quadrature demodulator for quadrature demodulation of an output signal of the
power amplifier into a fedback inphase component and a fedback quadrature phase component;
a first differential amplifier, connected upstream the quadrature modulator,
said first differential amplifier having a first input supplied by the inphase
component of the complex input signal and a second input supplied by the fedback
inphase component;
a second differential amplifier, connected upstream the quadrature modulator,
said second differential amplifier having a first input of the second differential
amplifier supplied by the quadrature phase component of the complex input signal
and a second input of the second differential amplifier supplied by the fedback
quadrature phase component; and
a phase modifier, which supplies to the quadrature demodulator an oscillator
sigal, shifted with regard to an oscillator signal supplied to the quadrature modulator
by the phase angle to be adjusted;
applying an input signal with a predetermined constant inphase component and
a predetermined constant quadrature phase component at each transmitting interval
with a closed feedback loop containing the quadrature modulator, the power amplifier,
the quadrature demodulator, the first differential amplifier and the second differential amplifier;
measuring the quadrature phase component and optionally the inphase component
at a first measuring point behind an output of the first differential amplifier
and a second measuring point behind an output of the second differential amplifier;
determining a phase correction value based on the measured quadrature phase component
and optionally the measured inphase component; and
correcting the currently set phase angle of the phase modifier by adding or subtracting
the determined phase correction value in a transmitting interruption interval,
wherein the quadrature phase component of the complex input signal being applied
has a value of zero, the measuring at the second measuring point behind the output
of the second differential amplifier takes place at a beginning of every transmitting
interval and the determining of the phase correction value comprises altering the
phase angle by a step width in a first direction if the measured quadrature phase
component is positive and altering the phase angle by a step width in an opposite
direction if the measured quadrature phase component is negative.
9. The method according to claim 8, wherein the inphase component is measured
at the first measuring point behind the output of the first differential amplifier.
10. The method according to claim 9, wherein the phase correction value (Δφ)
is determined by solving the following equation:
Δφ=arc tan (
VQM/VIM)-arc tan (
Q/I)
wherein V
QM is the measured quadrature phase component, V
IM is
the measured inphase component, Q is the predetermined quadrature phase component
and I is the predetermined inphase component.
11. The method according to claim 8, wherein the phase correction value (Δφ)
is determined by solving the following equation:
Δφ=arc tan (
VQM/VIM)-arc tan (
Q/I)
wherein V
QM is the measured quadrature phase component, V
IM is
the measured inphase component, Q is the predetermined quadrature phase component
and I is the predetermined inphase component.
12. The method according to claim 8, wherein the step width depends on an amount
of the measured quadrature component.
13. The method according to claim 8, wherein the phase angle is not altered if
the amount of the measured quadrature phase component is smaller than a predetermined
limit value.
14. The method according to claim 8, wherein prior to or concurrent with activating
the transmitting device, the phase angle of the phase modifier is preliminarily
adjusted such that an output power is measured at a power detector connected downstream
of the power amplifier and the phase angle is pre-adjusted such that a minimum
of the output power results.
15. A method for adjusting a phase angle of a phase modifier of a transmitting
device, wherein said method comprises:
providing the transmitting device comprising:
a quadrature modulator for quadrature modulation of an inphase component and
a quadrature phase component of a complex input signal;
a power amplifier, connected downstream of the quadrature modulator;
a quadrature demodulator for quadrature demodulation of an output signal of the
power amplifier into a fedback inphase component and a fedback quadrature phase component;
a first differential amplifier, connected upstream the quadrature modulator,
said first differential amplifier having a first input supplied by the inphase
component of the complex input signal and a second input supplied by the fedback
inphase component;
a second differential amplifier, connected upstream the quadrature modulator,
said second differential amplifier having a first input of the second differential
amplifier supplied by the quadrature phase component of the complex input signal
and a second input of the second differential amplifier supplied by the fedback
quadrature phase component; and
a phase modifier, which supplies to the quadrature demodulator an oscillator
signal, shifted with regard to an oscillator signal supplied to the quadrature
modulator by the phase angle to be adjusted;
applying an input signal with a predetermined constant inphase component and
a predetermined constant quadrature phase component at each transmitting interval
with a closed feedback loop containing the quadrature modulator, the power amplifier,
the quadrature demodulator, the first differential amplifier and the second differential amplifier;
measuring the quadrature phase component and optionally the inphase component
at a first measuring point behind an output of the first differential amplifier
and a second measuring point behind an output of the second differential amplifier;
determining a phase correction value based on the measured quadrature phase component
and optionally the measured inphase component; and
correcting the currently set phase angle of the phase modifier by adding or subtracting
the determined phase correction value in a transmitting interruption interval,
wherein prior to or concurrent with activating the transmitting device, the phase
angle of the phase modifier is preliminarily adjusted such that an output power
is measured at a power detector connected downstream of the power amplifier and
the phase angle is pre-adjusted such that a minimum of the output power results.
16. The method according to claim 15, wherein the phase angle is not altered
if the amount of the measured quadrature phase component is smaller than a predetermined
limit value.
17. The method according to claim 15, wherein the signal of the feedback loop
is damped during measurement of the output power.
18. The method according to claim 15, wherein the inphase component is measured
at the first measuring point behind the output of the first differential amplifier.
19. The method according to claim 18, wherein the phase angle is not altered
if the amount of the measured quadrature phase component is smaller than a predetermined
limit value.
20. The method according to claim 15, wherein the determining of the phase correction
value comprises altering the phrase angle by a step width in a first direction
if the measured quadrature phase component is positive and altering the phase angle
by a step width in an opposite direction if the measured quadrature inphase component
is negative.
Description
This application is a 371 of PCT/EP00/06078, filed Jun. 29, 2000, which claims
priority to Germany application No. 199 46 669.6, filed Sep. 29, 1999.
BACKGROUND OF THE INVENTION
The invention relates to a method for adjusting a phase angle of a phase modifier
of a transmitting device. The transmitting device comprises a quadrature modulator
and a power amplifier which is linearized via a so-called Cartesian feedback loop
with a quadrature demodulator.
EP 0 706 259 A1 discloses a transmitting device wherein a basic band input signal
is supplied to a quadrature modulator via two differential amplifiers. Said quadrature
modulator performs quadrature modulation of the inphase component and the quadrature
phase component of the complex input signal. Power amplification takes place in
a power amplifier connected downstream the quadrature modulator. To compensate
the non-linerarity of this power amplifier a feedback loop is provided, generally
designated as a Cartesian which separates the fedback signal into a fedback inphase
component and a fedback quadrature phase component. The fedback inphase component
is supplied, together with the inphase component of the input signal, to a first
differential amplifier, connected upstream the quadrature modulator. Correspondingly,
the fedback quadrature phase component is supplied, together with the quadrature
phase component of the input signal, to a second differential amplifier. In this
way the non-linearities of the power amplifier are compensated via the fedback signal.
In a transmitting device operating according to the Cartesian feedback method
it is particularly important that the fedback signal is input inphase. In order
to achieve this, the signal of a local oscillator, which is required for the quadrature
modulation and quadrature demodulation, is supplied to the quadrature demodulator
at a phase angle shifted with regard to the quadrature modulator. The phase shift
takes place in a phase modifier, the phase angle of which has to be adjusted. To
adjust the phase angle, in EP 0 706 259 A1 a test mode is proposed, in which the
feedback loop is interrupted at the output of the quadrature demodulator. A test
signal is applied to the input of the quadrature modulator and the output signal
of the quadrature demodulator is measured. With a predetermined input signal the
phase angle to be set can be calculated from the real part and the imaginary part
of the output signal of the quadrature demodulator.
Of disadvantage in the mode of operation proposed in EP 0 706 259 A1, however,
is that the feedback loop for determining the phase angle has to be opened each
time. This method may be suitable for adjusting the phase angle once on taking
into operation, but in the application of a transmitting device operating on the
Cartesian feedback principle in aeronautical radio, in particular with digital
aeronautical radio operating according to the VDL standard (VHL digital link) in
TDMA Simplex mode, there is a necessity to check and possibly re-adjust the phase
angle at each transmitting interval (transmitting burst). This cannot be done with
the method emerging from EP 0 706 259 A1, owing to the time-consuming separation
of the feedback loop and the complicated measurement process.
SUMMARY OF THE INVENTION
Therefore the object of the invention is to cite a method for adjusting
a phase angle of a phase modifier of a transmitting device with a power amplifier
which is linearized according to the Cartesian feedback principle, which enables
correction or re-setting of the phase angle at each transmitting interval.
The object is achieved by the characterising features of claim 1 in conjunction
with the generic features.
The invention is based on the awareness that, by applying an input signal with
a predetermined constant inphase component and a predetermined constant quadrature
phase component, a deviation of the phase angle can be relatively easily determined.
The feedback loop, consisting of quadrature modulator, power amplifier and quadrature
demodulator and the differential amplifiers, can therein remain closed. The method
can be carried out at each transmitting interval, as it is not time-consuming and
does not require separation of the feedback loop.
Claims 2 to 9 relate to advantageous further developments of
the method according to the invention.
Applying an input signal with predetermined inphase component (I=const.)
and without quadrature phase component (Q=0) and measuring the quadrature phase
component can advantageously take place at the output of the differential amplifier
at the beginning of every transmitting interval. During switching over from receive
mode to transmit mode it is in any case advantageous to apply, in addition, for
example, to three data symbols, a reference signal with a pure inphase component
without quadrature phase component at the beginning of the transmitting interval.
This reference signal can be used for determining the phase according to the invention
without taking extra time. With an input signal without quadrature phase component
(Q=0) ideally no voltage occurs at the output of the differential amplifier in
the quadrature phase control loop. If a voltage is nevertheless measured at this
measuring point this indicates a phase error, which can be corrected in the next
transmitting interruption interval or receiving interval.
The phase correction value can be immediately determined from the measured quadrature
phase component, optionally taking into account the additionally measured inphase
component, by an arcus tangens relation. The phase correction values assigned to
the measured values can be tabulated in a memory (look-up table) and read off immediately
without further calculation. An alternative possibility for determining the optimum
phase correction value consists in a trial and error method, in which the phase
angle is minimally altered experimentally during a receiving interval and in the
subsequent transmitting interval, by measuring the quadrature phase component with
the previously described reference signal, it is ascertained whether the newly
adjusted phase angle yields a better result. If this is the case, the phase angle
is further altered in this direction in the subsequent receiving interval. If the
newly set phase angle yields a worsening, in the subsequent receiving interval
the phase angle is adjusted back to the previously set value. Due to this fine
alignment, minimal phase fluctuations, resulting, for example, from a drift in
temperature, can be re-corrected while operation is running.
Before the transmitting device is taken into operation for the first time
it is advantageous to perform a preliminary adjustment of the phase angle in such
a way that a minimal output power results. For this case the maximum self-damping
of the system results, by contrast with the reverse case of maximum output power,
resulting in maximum positive feedback of the system. The signal of the feedback
loop is in this case damped.
A simplified embodiment example of the invention is described in greater detail
below with reference to the drawings.
BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS
FIG. 1 shows a block diagram of an embodiment example of a transmitting device,
suitable for the method according to the invention;
FIG. 2 shows a time-dependency diagram to explain the method according to the invention;
FIG. 3 shows a flow diagram to explain an embodiment example of the method according
to the invention and
FIG. 4 shows a diagram to explain the measuring of the phase correction angle.
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 shows a transmitting device
1 suitable for carrying out the method
according to the invention in a basic block diagram.
A digital signal processor (DSP)
2 generates a complex input signal for
a quadrature modulator
3, consisting of an inphase mixer
4, a quadrature
phase mixer
5 and a summer
6, as well as a phase modifier
7.
The complex input signal consists of an inphase component I and a quadrature phase
component Q, wherein the inphase component I is supplied to the inphase mixer
4
and the quadrature phase component Q is supplied to the quadrature phase mixer
5. The output signal of a local oscillator
8 is supplied to the phase
modifier
7, wherein the phase modifier
7 supplies this oscillator
signal to the inphase mixer
4 without phase shift and to the quadrature
phase mixer
5 at a phase shift of 90°.
Connected downstream the quadrature modulator
3 is a power amplifier
9 which supplies the quadrature-modulated signal, power-amplified corresponding
to the transmitting power of the transmitting device
1, to an antenna
13
via a circulator
10, a power detector
11 and a transmit-receive changeover
switch
12. In the embodiment example illustrated in FIG. 1 the digital signal
processor
2 acts simultaneously as control unit for the transmit-receive
changeover and triggers the transmit-receive changeover switch
12 in such
a way that the antenna
13 is connected to the power amplifier
9 in
transmit mode and to a receiver designated as an RX in receive mode. The circulator
10, connected to the terminal resistance
14, serves to avoid feedback
of possibly reflected transmitting power into the power amplifier
9.
In the signal path between the power amplifier
9 and the antenna
13
is a decoupler
15, which couples the output signal of the power amplifier
9 into a feedback loop
16. In the feedback loop
16 is a changeover
switch
17, via which an input
18 of a quadrature demodulator
19
can be optionally connected to the decoupler
15 or a terminal resistance
20. Between the decoupler
15 and the changeover switch
17
is a logarithmic power detector
39. The quadrature demodulator
19
consists of a signal divider
21, which divides the input signal equally
between an inphase mixer
22 and a quadrature phase mixer
23. Further
provided is a phase modifier
24, to which the output signal of the local
oscillator
8 is supplied via an adjustable phase modifier
25. Phase
modifier
24 operates like phase modifier
7 and supplies to the inphase
mixer
22 a non-phase-shifted oscillator signal and to the quadrature phase
mixer
23 an oscillator signal phase-shifted by 90°, wherein the oscillator
signal has previously been phase-shifted in total by a phase angle φ by the
phase modifier
25.
At the output of the inphase mixer
22 is a fedback inphase component I′
and at the output of the quadrature phase mixer
23 is a fedback quadrature
phase component Q′. The inphase component I of the input signal is passed
to the (+) input of a first differential amplifier
26, while the fedback
inphase component I′ is passed to the (-) input of the first differential
amplifier
26. Correspondingly the quadrature phase component Q of the input
signal is supplied to the (+) input of a second differential amplifier
27,
while the fedback quadrature phase component Q′ is supplied to the (-) input
of the second differential amplifier
27. By means of this feedback arrangement,
generally designated as Cartesian feedback, it is achieved that linearization errors
of the power amplifier
9 are compensated by the quadrature demodulator
19,
arranged in the feedback loop
16, and the differential amplifiers
26
and
27. It should therein be ensured, however, that the fedback signal I′,
Q′ is supplied to the differential amplifiers
26 and
27 with
a phase shift of 0° with regard to the input signal I, Q. The correct phase
position is set by the adjustable phase modifier
25, the phase angle φ
of which can be altered with the method according to the invention by the digital
signal processor via a control signal.
As both the quadrature modulator
3 and the quadrature demodulator
19
have a direct current offset (DC offset), this direct current offset has to be
correspondingly compensated. A third differential amplifier
28, arranged
between the inphase mixer
22 of the quadrature demodulator
19 and
the first amplifier
26, serves this purpose. A fourth differential amplifier
29 is arranged between the quadrature phase mixer
23 of the quadrature
demodulator
19 and the second differential amplifier
27. While the
fedback inphase component I′ is supplied to the (+) input of the third differential
amplifier
28, a first compensation voltage V
I1 is supplied to
the (-) input of the third differential amplifier
28, so the direct current
offset in the I′ component of the quadrature demodulator
19 is compensated
at the output of the third differential amplifier
28. Correspondingly the
fedback quadrature phase component Q′ is supplied to the fourth differential
amplifier
29 at its (+) input, while a fourth compensation voltage V
Q1
is supplied to its (-) input.
A fifth differential amplifier
30, to the (+) input of which the output
of the first differential amplifier
26 is supplied, while a third compensation
voltage V
I2 is supplied to its (-) input, serves to compensate the direct
current offset of the quadrature modulator
3. Further provided is a sixth
differential amplifier
31, the output of which is connected to the quadrature
phase mixer
5 of the quadrature modulator
3 and to the (+) input
of which the output of the second differential amplifier
27 is supplied.
A fourth compensation voltage V
Q2 is supplied to the (-) input of the
sixth differential amplifier
31. The compensation voltages V
I1,
V
Q1, V
I2, and V
Q2 are drawn in as controllable
voltage sources in FIG. 1 for better illustration, however these compensation voltages
are expediently generated internally in the digital signal processor
2.
With fast changeover between transmit mode and receive mode there is a problem,
where a feedback loop
16 according to the Cartesian feedback principle is
used, that the high frequency signal path of the loop, consisting of the quadrature
modulator
3, the power amplifier
9, the quadrature demodulator
19
and the differential amplifiers
26 and
27, has to be interrupted
during the changeover from transmit mode to receive mode, as the power amplifier
9 and the local oscillator
8 have to be switched off. When the power
amplifier
9 and the local oscillator
8 are switched on again and
the high frequency signal path is restored via the feedback loop
16, a switching
shock pulse is caused, as the voltages of the control system, in other words the
output voltages of the two differential amplifiers
26,
27, run to
the positive or negative control limit stop when the high frequency signal path
is open. This leads to an unacceptable sudden power variation to the maximum possible
transmitting power of the power amplifier
9. Therefore in FIG. 1, as well
as the high frequency signal path from the output of the differential amplifiers
26 and
27 via the quadrature modulator
3, the power amplifier
9 and the quadrature demodulator
19 to the (-) input of the differential
amplifiers
26 and
27, two direct DC signal paths
32 and
33
are to be provided which directly connect the output of the differential amplifier
26 or
27, assigned in each case, to the (-) input of the respective
differential amplifier
26 or
27. The direct DC signal paths
32
and
33 consist in the embodiment example illustrated in each case of a controllable
switch
34 or
35, which can be constructed, for example, as field-effect
transistors, and a resistance
36 or
37, connected in series. During
receive mode a constant OV potential can be maintained at the input and output
of the differential amplifier
26 and
27, so the changeover to transmit
mode takes place without shock pulses. The function of the low impedance resistances
51 and
52, arranged parallel to the resistances
36 and
37
and able to be connected to the switches
34 and
35 via a separate
switch position, is explained in more detail later.
FIG. 2 shows in a time-dependency diagram the sequence of the changeover from
receive mode to transmit mode. In the top partial diagram the output power TX is
represented logarithmically as a function of the time. Further illustrated in FIG.
2 and designated as RX is the signal of the latest possible receiving interval.
In the partial diagram below it the input signal I/Q is represented as a function
of the time. Below this is the signal "S/E" for actuating the transmit/receive
changeover switch
12 and the signal "DC loop" for actuating switches
34
and
35 in each case as a function of the time t. The signal "BIAS" designates
the supply voltage for the power amplifier
9, while the signal "LO level"
designates the level of the local oscillator
8.
As can be seen from FIG. 2, during the changeover from receive mode to transmit
mode the procedure is as follows:
First the level of the local oscillator
8 is increased. Then the supply
voltage (BIAS) for the power amplifier
9 is switched on and the switch
17
subsequently actuated, so the input of the quadrature demodulator
19 is
switched over to the decoupler
15. After the high frequency feedback loop
has thus been closed, switches
34 and
35 are opened by the signal
"DC loop" and the direct current paths
32 and
33 are thus interrupted.
Finally, by means of the signal "S/E" the transmit/receive changeover switch
12
is switched over to transmit mode. Subsequently the input signal I/Q can be supplied
to the quadrature modulator
3 via the (+) inputs of the differential amplifiers
26 and
27 and the output power TX thus successively increased (ramping).
In the time interval between times t
1 and t
2 an almost
constant
output signal is available. In the embodiment example an input signal I/Q is used
as reference signal between times t
1 and t
2, consisting of
a constant inphase component (I=const.) without quadrature phase component (Q=0).
This signal is applied as reference signal at the beginning of every transmitting
interval before transfer of the actual data for a period of preferably three data
symbols in the time interval between times t
1 and t
2. Simultaneously
at least the quadrature phase component V
QM is measured at measuring
point
53 in FIG. 1. Preferably the inphase component V
IM is also
measured at measuring point
61. Since a pure inphase component without quadrature
phase component is used as input signal, ideally, i.e. with correctly chosen phase
angle φ for the phase modifier
25, the measuring signal V
QM at
measuring point
53 is zero. If a deviating measuring voltage occurs this
indicates a phase error which is to be corrected.
The method according to the invention for adjusting the phase angle φ is
explained using FIG. 3. The method is divided into pre-adjusting of the phase angle
p, to be performed once when the transmitting device
1 is taken into operation
(method steps
40), re-setting the phase angle φ at each transmitting
interval (transmitting burst) (method steps
41) and optional fine re-setting
of the phase angle φ at each transmitting interval (method steps
42).
When the transmitting device
1 is taken into operation the phase angle
φ of the phase modifier
25 in the embodiment example illustrated in
FIG. 3 is pre-adjusted in such a way that the power P is measured as a function
of the phase angle φ by the logarithmic power detector
39 or by the
power detector
11. The phase angle φ is therein continually varied
in the range of 0° to 360°. Finally, the particular phase angle φ
in which the measurement resulted in the minimum power P
min is set.
This measuring principle is based on the assumption that for the phase angle φ
for which the minimum output power P
min results the feedback loop
16
is optimally negatively fed back. The thus pre-adjusted phase angle φ usually
offers a good starting point for the adjustment method to be described below, which
is performed at each transmitting interval. During this measurement of the output
power the signal of the feedback loop
16 is damped, in order to avoid too
great a positive feedback with a rough misadjustment of the phase angle φ,
with the danger of destroying the power amplifier
9. In the embodiment example
this damping is achieved in that switches
34 and
35 are switched
over to the low impedance resistances
51 and
52, in order to achieve
strong negative feedback of the differential amplifiers
26 and
27.
Alternatively series resistances, for example, could also be connected
in the feedback loop
16.
With the adjustment method according to the invention, as described, at the
beginning of every transmitting interval or transmitting burst a reference signal
with a pure inphase component (I=const.) without quadrature phase component (Q=0)
is applied for a period of preferably three data symbols and at least the quadrature
phase component (measuring signal V
QM) is measured at the output of
the second differential amplifier
27 at measuring point
53. As the
inphase component I is constant, it is sufficient to relate the measured quadrature
phase component V
QM to the input inphase component I and to use them
as an argument for the arcus tangens function, in order to obtain the phase correction
value Δφ. Accuracy of measurement can be increased in that the measured
quadrature phase component V
QM is related not to the predetermined inphase
component I at the input of the first differential amplifier
26, but to
the inphase component V
IM measured at the output of the first differential
amplifier
26. The corrected phase angle φ′ results from addition
of the phase correction angle Δφ to the previously adjusted phase angle
φ. The phase correction value Δφcan be read off in a stored table
as a function of the measured signal V
QM or V
QM and V
IM.
In the variant of the adjustment method described in FIG. 3 re-setting the phase
angle φ takes place by means of the arcus tangens function only until the
obtained phase correction value Δφ is greater than a predetermined
constant c. If the phase correction value Δφ is smaller than the limit
value c, a change is made to an iterative fine adjustment method
42. This
fine adjustment method
42 is based on a trial and error principle. Before
each transmitting burst the currently set phase angle φ is altered experimentally
by a step width Δφ
step and then at the beginning of the
transmitting burst the measuring voltage V
QM is measured at measuring
point
53, while at the input a pure inphase component I without quadrature
phase component applies. With a properly adjusted phase angle φ the measuring
voltage V
QM is ideally 0. If the amount |V
QM| of the measuring
voltage V
QM is reduced by the variation of the adjusted phase angle
φ, this newly adjusted phase angle φ′ is better than the previously
adjusted phase angle φ. The phase angle φ is optionally altered again
in this direction for the next transmitting burst, in order to test whether the
amount of the measuring voltage V
QM therein decreases even further.
The step width can optionally be varied as a function of the amount of the measuring
voltage V
QM. If the amount of the measuring voltage V
QM is
greater, however, the setting is put back to the previously set phase angle φ.
This method is then repeated in the opposite direction with reversed algebraic
sign of Δφ
step. If fine adjustment in the opposite direction
also does not result in an improvement, the previously set phase angle φ
is the best value and is left for a predetermined period. After a period after
which, for example, owing to a thermal drift, a shift in the phase angle φ
may have resulted, the method described above is repeated.
FIG. 4 shows the predetermined constant input signal (I, Q), consisting of the
inphase component I and the quadrature phase component Q, and the measuring signal
(V
IM, V
QM) measured at the output of the differential amplifiers
26 and
27, consisting of the measured inphase component VIM and the
measured quadrature phase component V
QM.
The predetermined desired phase angle φ
soll results therein
from the relation
##EQU1##
The measured actual phase angle φ
ist results from the relation
##EQU2##
The phase correction value Δφ results from the relation
##EQU3##
In the embodiment example described using FIG. 3 an input signal with a pure
inphase
component has been used, wherein the input quadrature phase component Q is zero,
so φ
sol1=0. As the preceding relation shows, other input signals
with other desired phase angles can also be used, however, wherein the use of the
desired phase angle φ
soll=0 is preferred owing to the resulting
simplification of the measuring process.
The invention is not restricted to the embodiment example illustrated. In particular
other algorithms than those illustrated in FIG. 3 can be used. The preliminary
adjustment of the phase angle φ illustrated in FIG. 3 can also be done in
other ways. Instead of an input signal with a pure inphase component any chosen
input signal with constant phase angle φ can be used.
*
