Switching power supply circuit Number:7,110,268 from the United States Patent and Trademark Office (PTO) owispatent

Title: Switching power supply circuit

Abstract: The present invention provides a power supply circuit performing constant voltage control by switching frequency control and achieving minimization of a necessary control range of the switching frequency control and implementation of a configuration ready for a wide range. A switching power supply circuit includes a switching circuit, a switching driving unit, an insulating converter transformer, a primary side series resonance circuit, a secondary side series resonance circuit, a secondary side DC output voltage production unit, a constant voltage control circuit, and a composite coupling coefficient setting mechanism.

Patent Number: 7,110,268 Issued on 09/19/2006 to Yasumura

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Inventors:	Yasumura; Masayuki (Kanagawa, JP)
Assignee:	Sony Corporation (JP)
Appl. No.:	11/141,346
Filed:	May 31, 2005

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Foreign Application Priority Data

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May 31, 2004 [JP]			P2004-162175
Jun 30, 2004 [JP]			P2004-194105
Sep 13, 2004 [JP]			P2004-265441
Sep 16, 2004 [JP]			P2004-269828
Feb 28, 2005 [JP]			P2005-054550

Current U.S. Class:	363/21.03
Current International Class:	H02M 3/335 (20060101)
Field of Search:	363/21.03,21.02,16

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References Cited [Referenced By]

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U.S. Patent Documents

	6483721		November 2002		Terashi
	6654259		November 2003		Koshita et al.

Foreign Patent Documents

	2003-235259		Aug., 2003		JP

Primary Examiner: Riley; Shawn
Attorney, Agent or Firm: Lerner,David,Littenberg,Krumholz & Mentlik,LLP

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Claims

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The invention claimed is:

1. A switching power supply circuit, comprising: a switching circuit including a switching device operable to perform a switching operation at a switching frequency based on a DC input voltage, the switching operation resulting in a switching output; a switching driving unit operable to drive the switching device to perform the switching operation; an insulating converter transformer having a core with a primary winding on a primary side and a secondary winding on a secondary side, the primary winding being supplied with the switching output of the switching operation, and the secondary winding having an alternating voltage induced therein by the primary winding, the core having a gap formed at a predetermined position between the primary side and the secondary side, the gap having a length selected to produce a predetermined coupling coefficient between the primary side and the secondary side; a primary side series resonance circuit including a leakage inductance component of the primary winding and a capacitance of a primary side series resonance capacitor connected in series with the primary winding for producing a predetermined primary side resonance frequency for making the switching circuit operate on a current resonance basis; a secondary side series resonance circuit including a leakage inductance component of the secondary winding and a capacitance of a secondary side series resonance capacitor connected in series with the secondary winding for producing a predetermined secondary side resonance frequency; the primary side series resonance circuit and the secondary side series resonant circuit forming an electromagnetic coupling type resonance circuit; a secondary side DC output voltage production unit operable to input a resonance output from the secondary side series resonance circuit and to perform a rectification operation on the input resonance output to produce a secondary side DC output voltage; a constant voltage control unit operable to control the switching driving unit in response to a level of the secondary side DC output voltage to adjust the switching frequency of the switching circuit to perform constant voltage control for the secondary side DC output voltage; and a composite coupling coefficient setting unit operable to set a composite coupling coefficient between the primary side and the secondary side of the insulating converter transformer so that the electromagnetic coupling type resonance circuit has a unimodal output characteristic with respect to an input of a frequency signal having the switching frequency.

2. The switching power supply circuit according to claim 1, wherein the secondary side resonance frequency is lower than the primary side resonance frequency.

3. The switching power supply circuit according to claim 1, wherein the secondary side resonance frequency is higher than the primary side resonance frequency.

4. The switching power supply circuit according to claim 1, wherein the secondary side resonance frequency has a predetermined value within a range between a lower limit frequency represented by a predetermined magnification value less than 1 and an upper limit frequency represented by a predetermined magnification value greater than 1 with respect to the primary side resonance frequency.

5. The switching power supply circuit according to claim 1, wherein the composite coupling coefficient setting unit includes the insulating converter transformer, and the length of the gap in the core of the insulating converter transformer is selected so that the composite coupling coefficient between the primary side and the secondary side of the insulting converter transformer is based on the coupling coefficient of the insulating converter transformer.

6. The switching power supply circuit according to claim 1, wherein the composite coupling coefficient setting unit includes: the insulating converter transformer, wherein the length of the gap in the core of the insulating converter transformer is selected so that the coupling coefficient of the insulating converter transformer has a predetermined value higher than the composite coupling coefficient; and an inductor having a predetermined inductance connected in series with the primary winding and/or the secondary winding of the insulating converter transformer.

7. The switching power supply circuit according to claim 1, wherein the composite coupling coefficient setting unit includes the insulating converter transformer, the insulating converter transformer including a magnetic path generating portion operable to equivalently form an inductor connected in series with the primary winding and/or the secondary winding of the insulating converter transformer.

8. The switching power supply circuit according to claim 7, wherein the core of the insulating converter transformer is an EE type core having a pair of outer magnetic legs and an inner magnetic leg, and the magnetic path generating portion includes a projection formed on at least one of the outer magnetic legs so as to project toward the inner magnetic leg.

9. The switching power supply circuit according to claim 1, wherein the secondary side DC output voltage producing unit includes a rectification device having on and off conditions, the switching power supply circuit further comprising a secondary side partial resonance capacitor having a capacitance that cooperates with the leakage inductance component of the secondary winding to form a partial resonance circuit, the partial resonance circuit being operable to perform a partial resonance operation when the rectification device is in the off condition.

10. The switching power supply circuit according to claim 1, wherein the switching circuit includes four switching devices connected in a full bridge connection.

11. The switching power supply circuit according to claim 1, further comprising a rectification smoothing circuit operable to input a commercial AC power supply voltage and to output the DC input voltage as a rectification smoothed voltage, the rectification smoothing circuit being a voltage doubler rectification circuit operable to form the rectification smoothed voltage at twice the voltage of the commercial AC power supply voltage.

12. The switching power supply circuit according to claim 1, wherein the secondary side DC output voltage producing unit includes a bridge rectification circuit including rectification devices connected in a bridge connection to produce a rectification output and a secondary side smoothing capacitor operable to smooth the rectification output, the bridge rectification circuit performing a full wave rectification operation.

13. The switching power supply circuit according to claim 1, wherein the secondary side DC output voltage producing unit is a voltage doubler rectification circuit operable to produce the secondary side DC output voltage at twice the voltage of the alternating voltage induced in the secondary winding.

14. The switching power supply circuit according to claim 13, further comprising a secondary side smoothing capacitor, wherein the voltage doubler rectification circuit is a voltage doubler half-wave rectification circuit formed so as to charge a rectification current within only ones of half cycles of the alternating voltage induced in the secondary winding into the secondary side smoothing capacitor to produce at least part of the secondary side DC output voltage as a voltage across the secondary side smoothing capacitor.

15. The switching power supply circuit according to claim 13, further comprising a secondary side smoothing capacitor, wherein the voltage doubler rectification circuit is a voltage doubler full-wave rectification circuit formed so as to charge a rectification current within both of half cycles of the alternating voltage induced in the secondary winding into the secondary side smoothing capacitor to produce at least part of the secondary side DC output voltage as a voltage across the secondary side smoothing capacitor.

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Description

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CROSS-REFERENCE TO RELATED APPLICATIONS

The present application claims priority from Japanese Application Nos. 2004-162175 filed May 31, 2004, 2004-194105 filed Jun. 30, 2004, 2004-265441 filed Sep. 13, 2004, 2004-269828 filed Sep. 16, 2004, and 2005-054550 filed Feb. 28, 2005, the disclosures of which are hereby incorporated by reference herein.

BACKGROUND OF THE INVENTION

This invention relates to a switching power supply circuit to be used as a power supply for various electronic apparatus.

Various power supply circuits having a resonance type converter on the primary side have been proposed by the assignee of the present application. One such power supply circuit is disclosed, for example, in Japanese Patent Laid-Open No. 2003-235259 (hereinafter referred to as Patent Document 1).

FIG. 44 shows the switching power supply circuit disclosed in Patent Document 1 which includes a resonance type converter. Referring to FIG. 44, the power supply circuit shown includes a switching converter. The switching converter is configured such that a partial voltage resonance circuit for performing a voltage resonance operation only upon turn-off during switching is combined with a separately excited current resonance type converter of a half bridge coupling scheme.

The switching power supply circuit shown in FIG. 44 is provided, for example, as a power supply of a printer apparatus. In the printer apparatus, for example, the load power exhibits a variation over a comparatively wide range from approximately 100 W or more to no load.

In the power supply circuit shown in FIG. 44, a common mode noise filter formed from two filter capacitors CL and a common mode choke coil CMC is connected to a commercial AC power supply AC.

Further, as a rectification smoothing circuit for producing a DC input voltage from the commercial AC power supply AC, a full wave rectification circuit formed from a bridge rectification circuit Di and a smoothing capacitor Ci is provided at the stage following the common mode noise filter.

The rectification output of the bridge rectification circuit Di is charged into the smoothing capacitor Ci, and as a result, a rectification smoothed voltage Ei (DC input voltage) having a level equal to that of an AC input voltage VAC is obtained across the smoothing capacitor Ci.

The current resonance type converter for receiving and switching the DC input voltage includes a switching circuit system wherein two switching devices Q1 and Q2, each formed from a MOS-FET, are connected to each other in a half bridge coupling scheme as seen in FIG. 44. Damper diodes DD1 and DD2 each formed from a body diode are individually connected in parallel to each other and in such directions as seen in FIG. 44 between the drain-source of the switching devices Q1 and Q2, respectively.

Further, a partial resonance capacitor Cp is connected in parallel between the drain-source of the switching device Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed from the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of a primary winding N1. The partial voltage resonance circuit performs a partial voltage resonance operation wherein it voltage resonates only when the switching devices Q1 and Q2 are turned off.

In the power supply circuit, in order to switching drive the switching devices Q1 and Q2, an oscillation and driving circuit 2 formed from, for example, a general-purpose IC is provided. The oscillation and driving circuit 2 includes an oscillation circuit and a driving circuit, and applies a driving signal (gate voltage) having a required frequency to the gates of the switching devices Q1 and Q2. Consequently, the switching devices Q1 and Q2 perform a switching operation such that they are turned on and off alternately at a required switching frequency.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching devices Q1 and Q2 to the secondary side.

In this instance, the primary winding N1 of the insulating converter transformer PIT is connected at one end thereof to a node (switching output point) between the source of the switching device Q1 and the drain of the switching device Q2 through a series connection with a primary side series resonance capacitor C1 so that the switching output can be obtained from the node. The primary winding N1 is connected at the other end thereof to the primary side ground as seen in FIG. 44.

In this instance, the series resonance capacitor C1 and the primary winding N1 are connected in series to each other. Thus, a primary side series resonance circuit for making the operation of the switching converter that of a current resonance type is formed from the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the primary winding N1 (series resonance winding) of the insulating converter transformer PIT.

From the description above, resonance operation of the current resonance type by the primary side series resonance circuit (L1 C1) and partial voltage resonance operation by the partial voltage resonance circuit (Cp//L1) described hereinabove can be obtained by the primary side switching converter shown in FIG. 44.

In particular, the power supply circuit shown in FIG. 44 adopts a form in which the resonance circuit for making the operation of the primary side switching converter that of the resonance type is combined with another resonance circuit. A switching converter of the type just described is hereinafter referred to as a composite resonance type converter.

While a description with reference to the drawings is omitted here, the insulating converter transformer PIT described above is structured such that it includes an EE type core formed by combining E type cores made of, for example, a ferrite material with each other. Further, the primary winding N1 and a secondary winding N2 are wound on an inner magnetic leg of the EE type core at winding portions provided divisionally for the primary and secondary sides.

Further, a gap of 1.5 mm or less is formed in the inner magnetic leg of the EE type core of the insulating converter transformer PIT. Consequently, a coupling coefficient of 0.75 or more is obtained between the primary winding N1 and the secondary winding N2.

For the secondary winding N2 of the insulating converter transformer PIT, a full wave rectification circuit is provided. The full wave rectification circuit is formed from a bridge rectification circuit formed from rectification diodes Do1 to Do4, and a smoothing capacitor Co.

Consequently, a secondary side DC output voltage Eo, which is a DC voltage having a level equal to that of the alternating voltage induced in the secondary winding N2, can be obtained as a voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied as a main DC power supply to a main load (not shown), and is also branched and input as a detection voltage for constant voltage control to the control circuit 1.

The control circuit 1 outputs a control signal to the oscillation and driving circuit 2 in the form of a current or a voltage whose level is adjusted corresponding to that of the secondary side DC output voltage Eo.

The frequency of an oscillation signal produced by the oscillation circuit in the oscillation and driving circuit 2 is adjusted based on the control signal input from the control circuit 1 to adjust the frequency of the switching driving signal to be applied to the gates of the switching devices Q1 and Q2. Consequently, the switching frequency is adjusted. Since the switching frequency of the switching devices Q1 and Q2 is adjustably controlled in response to the level of the secondary side DC output voltage Eo in this manner, the resonance impedance of the primary side series resonance circuit is varied and also the energy to be transferred from the primary winding N1, which forms the primary side series resonance circuit, to the secondary side is varied. Further, at this time, the level of the secondary side DC output voltage Eo is also adjustably controlled. Consequently, constant voltage control for the secondary side DC output voltage Eo can be implemented.

It is to be noted that such constant voltage control for adjustably controlling the switching frequency to achieve stabilization as described above is hereinafter referred to as the "switching frequency controlling method".

FIG. 46 is a waveform diagram illustrating the operation of part of the power supply circuit shown in FIG. 44. Referring to FIG. 46, waveforms on the left side indicate the operation when the load power Po is Po=150 W, and waveforms on the right side indicate the operation of the same portions when the load power Po is Po=25 W. As an input voltage condition, the AC input voltage VAC is fixed at VAC=100 V.

Referring to FIG. 46, the voltage V1 of a rectangular wave is a voltage across the switching device Q2 and indicates on and off timings of the switching device Q2. The period that the voltage V1 is at the 0 level is an on period in which the switching device Q2 conducts. Within the on period, switching current IQ2 having the waveform illustrated in FIG. 46 is supplied to the switching circuit system formed from the switching device Q2 and the clamp diode DD2. Further, the period that the voltage V1 is clamped at the level of a rectification smoothed voltage Ei is a period in which the switching device Q2 is off, and the switching current IQ2 has the zero level as seen in FIG. 46.

Further, though not shown, the voltage obtained across the other switching device Q1 and the switching current flowing to the switching circuit (Q1, DD1) have waveforms shifted by 180.degree. in phase from the waveforms of the voltage V1 and the switching current IQ2, respectively. In short, as described above, the switching devices Q1 and Q2 perform the switching operation such that they are turned on and off alternately.

Further, the switching currents flowing to the switching circuits (Q1, DD1 and Q2, DD2) are composed to obtain a current having the waveform shown in FIG. 46, and the resulting current is used as the primary side series resonance current Io to flow to the primary side series resonance circuit (C1 N1(L1)).

Further, it can be recognized, for example, from a comparison between the waveform of the voltage V1 shown in FIG. 46 when the load power Po=150 W and the waveform of the voltage V1 shown in FIG. 46 when the load power Po=25 W that the switching frequency is controlled. The switching frequency on the primary side, when the load to the secondary side DC output voltage Eo is heavy (Po=150 W), is lower than that when the load to the secondary side DC output voltage Eo is light (Po=25 W). In particular, the switching frequency decreases in response to a decrease in the level of the secondary side DC output voltage Eo as the load becomes heavier, but the switching frequency increases in response to an increase in the level of the secondary side DC output voltage Eo as the load becomes lighter. This indicates the fact that a constant voltage controlling operation by upper side control is performed as a switching frequency controlling method.

Further, since the operation on the primary side described above is obtained, an alternating voltage V2 having the waveform shown in FIG. 46 is induced in the secondary winding N2 of the insulating converter transformer PIT. Then, within the period of any one of the half cycles in which the alternating voltage V2 has a positive polarity, the rectification diodes [Do1, Do4] on the secondary side conduct to allow rectification current ID1 to flow in the waveform and at the timing shown in FIG. 46. Further, within the period of the other one of the half cycles in which the alternating voltage V2 has a negative polarity, the rectification diodes [Do2, Do3] on the secondary side conduct to allow rectification current ID3 to flow in the waveform and at the timing shown in FIG. 46. Further, as seen in FIG. 46, the rectification currents ID1 and ID3 are composed to form secondary winding current I2 to flow to the secondary winding N2.

FIG. 47 illustrates an AC.fwdarw.DC power conversion efficiency and a characteristic of the switching frequency of the power supply circuit shown in FIG. 44 with respect to the load variation under the input voltage condition of the AC input voltage VAC=100 V.

The switching frequency fs has a characteristic that it decreases as the load increases because the constant voltage controlling operation is performed. However, the characteristic just described is not a linear variation characteristic with respect to the load variation. For example, within a range from the load power Po=approximately 25 W to Po=0 W or less, the tendency is exhibited that the switching frequency fs increases steeply.

Meanwhile, the AC.fwdarw.DC power conversion efficiency (.eta.AC.fwdarw.DC) has a tendency that it increases as the load power Po increases, and when the load power Po=150 W, an .eta.AC.fwdarw.DC power conversion efficiency of approximately 91.0% is obtained.

It is to be noted that, in order to obtain the experimental results described with reference to FIGS. 46 and 47, the components shown in FIG. 44 are set as mentioned just below. Insulating converter transformer PIT (EER-35 type ferrite core, gap length=1.4 mm, coupling coefficient k=0.75) Primary winding N1=35 T (turns), secondary winding N2=50 T Primary side series resonance capacitor C1=0.039 .mu.F Partial resonance capacitor Cp=330 pF

Another example of the switching power supply circuit is shown in the circuit diagram of FIG. 45. It is to be noted that, in FIG. 45, like elements to those of FIG. 44 are denoted by like reference characters, and a description of them is omitted herein to avoid redundancy.

The secondary side rectification circuit of the power supply circuit shown in FIG. 45 includes a full wave rectification circuit. In particular, a center tap is provided for the secondary winding N2 such that the secondary winding N2 is divided into secondary winding sections N2A and N2B. In this instance, the secondary winding sections N2A and N2B are formed from numbers of turns equal to each other. Further, the center tap is grounded to the secondary side ground. Furthermore, the rectification diodes Do1 and Do2 and the secondary side smoothing capacitor Co are connected to the secondary winding N2. By the full wave rectification circuit, the secondary side DC output voltage Eo can be obtained as a voltage across the smoothing capacitor Co.

A power supply circuit having the configuration described above may be provided as the power supply of a plasma display apparatus. In the plasma display apparatus, the load power Po varies over a comparatively wide range, for example, from Po=100 W or more to no load. Further, a secondary side DC output voltage having, for examples a comparatively high level of 200 V or more is required.

When an experiment regarding the power supply circuit shown in FIG. 45 was performed, results of operation and a characteristic substantially equal to those illustrated in FIGS. 46 and 47 were obtained.

It is to be noted that, when the experiment was performed, the components of the circuit shown in FIG. 45 were set as given below. Insulating converter transformer PIT (EER-35 type ferrite core, gap length=1.4 mm, coupling coefficient k=0.75) Primary winding N1=35 T (turns), secondary winding N2=secondary winding section N2A+secondary winding N2B=50 T+50 T=100 T Primary side series resonance capacitor C1=0.039 .mu.F Partial resonance capacitor Cp=330 pF

As described above, the power supply circuit shown in FIG. 45 is provided as a power supply of a plasma display apparatus and is configured so that the secondary side DC output voltage Eo obtained has a comparatively high level. In order to cope with this, in the circuit shown in FIG. 45, the secondary side rectification circuit is formed as a full wave rectification circuit, and the number of turns of the secondary winding N2 is suitably increased to 100 T.

Incidentally, where the configuration as a resonance type converter for implementing stabilization of the secondary side DC output voltage by the switching frequency controlling method is adopted as in the case of the power supply circuit shown in FIG. 44 (FIG. 45), the adjustable controlling range of the switching frequency for stabilization is a comparatively wide range.

This is described with reference to FIG. 48. FIG. 48 illustrates a constant voltage controlling characteristic of the power supply circuit shown in FIG. 44 (FIG. 45) in the form of a relationship between the level of the switching frequency fs and the level of the secondary side DC output voltage Eo.

It is to be noted that, in the description given with reference to FIG. 48, it is a premise that the power supply circuit in FIG. 44 (FIG. 45) adopts upper side control as the switching frequency controlling method. Upper side control as used herein is a controlling method for adjustably controlling the switching frequency within a frequency range higher than a resonance frequency fo1 of the primary side series resonance circuit and utilizing the variation of the resonance impedance caused by the adjustment control to control the level of the secondary side DC output voltage Eo.

Generally, the resonance impedance of the series resonance circuit is lowest at the resonance frequency fo1. Consequently, as a relationship between the secondary side DC output voltage Eo and the switching frequency fs in upper side control, the level of the secondary DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo1, but decreases as the switching frequency fs moves away from the resonance frequency fo1.

Accordingly, as seen in FIG. 48, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs in a condition that the load power Po is constant exhibits a quadratic curve variation. In particular, the level of the secondary side DC output voltage Eo exhibits a peak when the switching frequency fs is equal to the resonance frequency fo1 of the primary side series resonance circuit, and decreases as the switching frequency fs moves away from the resonance frequency fo1.

Further, the level of the secondary side DC output voltage Eo corresponding to the switching frequency fs in the same condition as that described above exhibits a characteristic that it shifts such that the level at the maximum load power Pomax is less by a predetermined amount than the level at the minimum load power Pomin. In particular, where it is considered that the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the load condition becomes heavier.

If an attempt is made to stabilize the secondary side DC output voltage Eo by upper side control so that Eo=tg may be satisfied where such a characteristic as just described is exhibited, then the adjustment range (necessary control range) of the switching frequency necessary for the power supply circuit shown in FIG. 44 (FIG. 45) is a range indicated by the reference character .DELTA. fs.

Actually, the power supply circuit shown in FIG. 44 performs the constant voltage control based on the switching frequency controlling method so that the secondary side DC output voltage Eo is stabilized at 135 V so as to cope with the input variation range and the load conditions. The variation range is from the AC input voltage VAC=85 V to 120 V of the AC 100 V type. The load conditions are the maximum load power Pomax=150 W and the minimum load power Pomin=0 W (no load) to the secondary side DC output voltage Eo, which is a main DC power supply.

In this instance, the variation range of the switching frequency fs that is varied for the constant voltage control by the power supply circuit shown in FIG. 44 is fs=80 kHz to 200 kHz or more, and also the range .DELTA. fs is a correspondingly wide range of 120 kHz or more.

Further, the power supply circuit shown in FIG. 45 performs the constant voltage control so that the secondary side DC output voltage Eo is stabilized at the rated level of approximately 200 V. Therefore, similar to the power supply circuit shown in FIG. 44, the range .DELTA. fs of the power supply circuit shown in FIG. 45 is a correspondingly wide range.

As one of the power supply circuits, a power supply circuit ready for a wide range is known, which is configured so as to operate with an AC input voltage range, for example, from approximately AC 85 V to 288 V. It can be applied both in an area in which the AC input voltage of the AC 100 V type is used, such as, for example, Japan, U.S.A. and so forth, and in another area in which the AC input voltage of the AC 200 V type is used, such as, for example, Europe.

Thus, it is considered here that the power supply circuit shown in FIG. 44 (FIG. 45) is configured as a power supply circuit ready for a wide range as described above.

As described above, where the power supply circuit is ready for a wide range, it is ready for an AC input voltage range, for example, from AC 85 V to 288 V. Accordingly, the variation range of the level of the secondary side DC output voltage Eo increases when compared with that in an alternative case in which the power supply circuit is ready for a single range of, for example, only the AC 100 V type or only the AC 200 V type. In order to perform the constant voltage control for the secondary side DC output voltage Eo whose variation range is expanded in correspondence to such an AC input voltage range as described above, switching frequency control over a still wider range is required. For example, in the power supply circuit shown in FIG. 44 (FIG. 45), it is necessary to expand the controlling range of the switching frequency fs to approximately 80 kHz to 500 kHz.

However, the upper limit to the driving frequency with which an IC (oscillation driving circuit 2) for driving an actual switching device can cope is approximately 200 kHz. Further, even if a switching driving IC that can be driven with such a high frequency as described above is configured and mounted, when the switching device is driven with such a high frequency as described above, the power conversion efficiency drops remarkably. Therefore, it is difficult to practically use the switching driving IC described above as an actual power supply circuit. Incidentally, the upper limit to the AC input voltage VAC which can be stabilized, for example, by the power supply circuit shown in FIG. 44 (FIG. 45), is approximately 100 V.

Therefore, in order to configure a switching power supply circuit using switching frequency control for stabilization as a switching power supply circuit ready for a wide range, it is known to adopt, for example, such a configuration as described below.

In particular, a rectification circuit system receiving a commercial AC power supply to produce the DC input voltage (Ei) is provided with a function of performing a changeover between a voltage doubler rectification circuit and a full wave rectification circuit in response to an input of commercial AC power supply of the AC 100 V type and the AC 200 V type.

In this instance, the circuit is configured such that the level of the commercial AC power supply is detected and the circuit connection of the rectification circuit system is changed over by a switch using an electromagnetic relay such that a voltage doubler rectification circuit or a full wave rectification circuit is formed in response to the detected level.

However, in such a configuration for the changeover of the rectification circuit system as described above, a number of electromagnetic relays are required as described above. Further, at least a pair of smoothing capacitors must be provided in order to form a voltage doubler rectification circuit. Therefore, the number of parts increases and this increases the cost. Also, the mounting area of a power supply circuit board is expanded to increase the size of the circuit. In particular, since the smoothing capacitors and the electromagnetic relays are large-size parts from among those parts forming the power supply circuit, the size of the board becomes rather large.

Where the configuration for changeover between full wave rectification operation and voltage doubler rectification operation is applied, if, when the commercial AC power supply of the AC 200 V type is input, the level of the AC input voltage becomes lower than that corresponding to the AC 200 type because an instantaneous service interruption occurs or the AC input voltage decreases to a voltage level lower than the rated voltage or the like, then a malfunction may occur in which it is detected that the commercial AC power supply input is the AC 100 V type and a changeover from the full wave rectification circuit to the voltage doubler rectification circuit is performed. If such a malfunction occurs, then the voltage doubler rectification will be performed for an AC input voltage having the level of the AC 200 V type. Therefore, there is the possibility that, for example, the switching devices Q1, Q2 and so forth may be broken by being subjected to a voltage higher than that which they can withstand.

Therefore, as an actual circuit, in order to prevent the occurrence of such a malfunction as described above, a configuration is adopted in which not only is the DC input voltage of the switching converter on the main power supply side detected, but the DC input voltage of the converter circuit on the standby power supply side is also detected. Consequently, a member for detecting the converter circuit on the standby power supply side must be added, and an increase of the cost described above and an increase of the size of the circuit board result.

Further, the DC input voltage of the converter on the standby power supply side is detected in order to prevent a malfunction. This signifies that only an electronic apparatus which includes not only a main power supply but also a standby power supply can actually use a power supply circuit which includes a circuit for changing over the rectification operation and which is ready for a wide range. In other words, the type of electronic apparatus in which the power supply can be incorporated is limited to that which includes a standby power supply, and the range of utilization becomes much narrower.

Further, as a configuration ready for a wide range, a configuration is also known in which the form of the current resonance converter on the primary side is changed over between that of a half bridge connection and that of a full bridge connection in response to an input of a commercial power supply of the A 100 V type/AC 200 V type.

With the configuration just described, even if the AC input voltage of the AC 200 V type drops to the level of the AC 100 V type, for example, as a result of such instantaneous interruption as described above or the like to cause a malfunction, the switching operation is only changed over from a half bridge operation to a full bridge operation. As a result, a situation does not arise in which a voltage higher than the withstanding voltage is applied to the switching devices. Therefore, the DC input voltage on the standby power supply side need not be detected. Consequently, the configuration can be applied to an electronic apparatus that does not include a standby power supply. Further, since a changeover of the commercial power supply line is not involved, a changeover of the circuit formation by a semiconductor switch is possible. Therefore, a large-size switching member such as an electromagnetic relay need not be provided.

However, with the configuration described above, at least four switching devices must be provided in order to form a full bridge connection in response to the AC 100 V type. In particular, in comparison with the configuration of a converter using only a half bridge connection, which can be formed from two switching devices, an additional two switching devices must be added.

Further, in the configuration described, four switching devices perform the switching operation in the full bridge operation, and three switching devices perform the switching operation even in the half bridge operation. While the resonance converter produces low switching noise, the disadvantage in regard to switching noise increases as the number of switching devices performing switching operations in such a manner as described above increases.

Also, where any one of the configurations described above is adopted as a configuration ready for a wide range in such a manner as described above, when compared with an alternative configuration ready for a single range, an increase in the circuit scale and an increase in the cost caused by an increase in the number of parts cannot be avoided. Further, intrinsic problems which do not appear with the configuration ready for a single range, such as a limitation on the range of apparatuses which can be utilized and an increase in the switching noise and so forth occur with the former configuration and the latter configuration, respectively.

Further, where the control range of the switching frequency is a suitably wide range as in the case of the power supply circuit shown in FIG. 44 (FIG. 45), a problem also occurs that a high-speed response characteristic in the stabilization of the secondary side DC output voltage Eo decreases.

Some electronic apparatus involve varying operations such that the load condition changes over instantaneously, for example, between a state in which the load has a maximum level and another state in which the load is substantially zero. A load exhibiting such a load variation as just described is also called a switching load. The power supply circuit to be incorporated in such an apparatus as just described must be configured so that the secondary side DC output voltage is appropriately stabilized against a load variation such as that of a switching load.

However, as described above with reference to FIG. 48, where the switching frequency has a characteristic of a wide control range, a comparatively long time is required to adjust the switching frequency with which the secondary side DC output voltage is provided to the required level in response to a load variation such as that of the switching load described above. In short, an undesirable result is obtained as the response characteristic of the constant voltage control.

It can be recognized that the power supply circuit shown in FIG. 44 (FIG. 45) is particularly disadvantageous in the constant voltage control response characteristic to a switching load as described above. In the switching frequency characteristic according to the constant voltage control, the switching frequency varies by a great amount within the load range from the load power Po=approximately 25 W to 0 W as seen in FIG. 47.

SUMMARY OF THE INVENTION

According to the present invention, there is provided a switching power supply circuit including a switching circuit, a switching driving unit, an insulating converter transformer, a primary side series resonance circuit, a secondary side series resonance circuit, a secondary side DC output voltage production unit, a constant voltage control unit, and a composite coupling coefficient setting unit. The switching circuit includes a switching device operable to perform a switching operation at a switching frequency based on a DC input voltage. The switching driving unit drives the switching device to perform the switching operation. The insulating converter transformer has a core with a primary winding on a primary side and a secondary winding on a secondary side, the primary winding being supplied with the switching output of the switching operation, and the secondary winding having an alternating voltage induced therein by the primary winding. The core has a gap formed at a predetermined position between the primary side and the secondary side, the gap having a length selected to produce a predetermined coupling coefficient between the primary side and secondary side. The primary side series resonance circuit includes a leakage inductance component of the primary winding and a capacitance of a primary side series resonance capacitor connected in series with the primary winding for producing a predetermined primary side resonance frequency for making the switching circuit operate on a current resonance basis. The secondary side series resonance circuit includes a leakage inductance component of the secondary winding and a capacitance of a secondary side series resonance capacitor connected in series with the secondary winding for producing a predetermined secondary side resonance frequency. The primary side series resonance circuit and the secondary side series resonance circuit form an electromagnetic coupling type resonance circuit. The secondary side DC output voltage production unit is operable to input a resonance output from the secondary side series resonance circuit and to perform a rectification operation on the input resonance output to produce a secondary side DC output voltage. The constant voltage control unit is operable to control the switching driving unit in response to a level of the secondary side DC output voltage to adjust the switching frequency of the switching circuit to perform constant voltage control for the secondary side DC output voltage. The composite coupling coefficient setting unit is operable to set a composite coupling coefficient between the primary side and the secondary side of the insulating converter transformer so that the electromagnetic coupling type resonance circuit has a unimodal output characteristic with respect to an input of a frequency signal having the switching frequency.

The switching power supply circuit adopts the basic configuration of a current resonance type converter including a primary side series resonance circuit, and also forms a series resonance circuit on the secondary side from the secondary winding and a secondary side series resonance capacitor.

Where the configuration described is adopted, the switching power supply circuit of the present invention includes a coupling type resonance capacity formed by the magnetic coupling of the insulating converter transformer. Further, the composite coupling coefficient between the primary side and the secondary side of the insulating converter transformer in the power supply circuit is set so that a unimodal output characteristic is obtained with respect to an input of a frequency signal (switching output) having the switching frequency. Since the unimodal characteristic is obtained in this manner, the variation range or necessary control range of the switching frequency required for stabilization can be reduced when compared with an alternative case wherein a series resonance circuit is formed only on the primary side.

In this manner, with the switching power supply circuit, the variable control range, that is, the necessary control range, of the switching frequency necessary for the constant voltage control is reduced.

Consequently, a resonance type converter ready for a wide range only by the switching frequency control can be readily obtained. Where a wide range can be achieved with only the switching frequency control, the necessity, for example, to change over a rectification circuit system in response to the level of a commercial AC power supply or to adopt a configuration for changing over a switching circuit system between a half bridge connection and a full bridge connection is eliminated. Consequently, advantages can be realized, including a reduction in the number of circuit components and a reduction in the board area, that the range of application of the power supply circuit to electronic apparatus is expanded, and that the switching power supply circuit is tough against switching noise.

As a basic configuration for achieving such advantages of the present invention as described above, it is only necessary to add a secondary side series resonance capacitor to the configuration of the current resonance type converter including the primary side series resonance circuit, and to adopt a configuration for the provision of an inductor connected in series with the primary winding and/or the secondary winding. Therefore, the need to increase the number of parts or to alter the parts is very small.

Further, where the necessary control range of the switching frequency is reduced as described above, the reliability of the constant voltage control is improved. This makes it possible to perform constant voltage control with a higher reliability than ever for a load variation called a switching load which varies in a switching manner between a maximum load condition and a no-load condition. This enhances the reliability of an apparatus in which the switching power supply circuit is incorporated.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other objects of the invention will be seen by reference to the description, taken in connection with the accompanying drawings, in which:

FIG. 1 is a circuit diagram showing an example of the configuration of a power supply circuit according a first embodiment of the present invention;

FIG. 2 is a sectional view showing an example of the structure of an insulating converter transformer provided in the switching power supply circuit of the first embodiment;

FIG. 3 is a waveform diagram illustrating the operation, in a maximum load power condition, of several components of the power supply circuit of the first embodiment;

FIG. 4 is a waveform diagram illustrating the operation, in a light load condition, of the several components of the power supply circuit of the first embodiment;

FIG. 5 is a circuit diagram showing an equivalent circuit to the power supply circuit of the first embodiment, which is viewed as an electromagnetic connection type resonance circuit;

FIG. 6 is a waveform diagram illustrating a constant voltage controlling characteristic of the power supply circuit of the first embodiment;

FIG. 7 is a waveform diagram illustrating a switching frequency control range or necessary control range with respect to an AC input voltage condition and a load variation as a constant voltage controlling operation of the power supply circuit of the first embodiment;

FIG. 8 is a graph illustrating the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation of the power supply circuit of the first embodiment;

FIG. 9 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the first embodiment;

FIG. 10 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the first embodiment;

FIG. 11 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second embodiment of the present invention;

FIG. 12 is a waveform diagram illustrating the operation, in a maximum load power condition, of several components of the power supply circuit of the second embodiment;

FIG. 13 is a waveform diagram illustrating the operation, in a light load condition, of several components of the power supply circuit of the second embodiment;

FIG. 14 is a graph illustrating the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation of the power supply circuit of the second embodiment;

FIG. 15 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the second embodiment;

FIG. 16 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the second embodiment;

FIG. 17 is a circuit diagram showing an example of the configuration of a power supply circuit according to a third embodiment of the present invention;

FIG. 18 is a waveform diagram illustrating the operation, in a maximum load power condition, of several components of the power supply circuit of the third embodiment;

FIG. 19 is a waveform diagram illustrating the operation, in a light load condition, of several components of the power supply circuit of the third embodiment;

FIG. 20 is a graph illustrating the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation of the power supply circuit of the third embodiment;

FIG. 21 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the third embodiment;

FIG. 22 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the third embodiment;

FIG. 23 is a circuit diagram showing an example of the configuration of a power supply circuit according to a fourth embodiment of the present invention;

FIG. 24 is a waveform diagram illustrating the operation of several components of the power supply circuit of the fourth embodiment;

FIG. 25 is a graph illustrating the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation regarding the power supply circuit of the fourth embodiment;

FIG. 26 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the fourth and seventh embodiments;

FIG. 27 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the fourth and seventh embodiments;

FIG. 28 is a circuit diagram showing an example of the configuration of a power supply circuit according to a fifth embodiment of the present invention;

FIG. 29 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the fifth and eighth embodiments;

FIG. 30 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the fifth and eighth embodiments;

FIG. 31 is a circuit diagram showing an example of the configuration of a power supply circuit according to a sixth embodiment of the present invention;

FIG. 32 is a circuit diagram showing an example of the configuration of a power supply circuit according to a first modification of the sixth and ninth embodiments;

FIG. 33 is a circuit diagram showing an example of the configuration of a power supply circuit according to a second modification of the sixth and ninth embodiments;

FIG. 34 is a circuit diagram showing an example of the configuration of a power supply circuit according to a seventh embodiment of the present invention;

FIG. 35 is a waveform diagram illustrating the operation of several components of the power supply circuit of the seventh embodiment;

FIG. 36 is a graph illustrating the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation regarding the power supply circuit of the seventh embodiment;

FIG. 37 is a circuit diagram showing an example of the configuration of a power supply circuit according to an eighth embodiment of the present invention;

FIG. 38 is a circuit diagram showing an example of the configuration of a power supply circuit according to a ninth embodiment of the present invention;

FIG. 39 is a sectional view showing an example of the structure of an insulating converter transformer used in a tenth embodiment of the present invention;

FIGS. 40A and 40B are sectional views illustrating magnetic paths of the insulating converter transformer used in the tenth embodiment;

FIGS. 41 to 43 are circuit diagrams showing equivalent circuits used in the power supply circuit according to the tenth embodiment;

FIGS. 44 and 45 are circuit diagrams showing different examples of the configuration of a power supply circuit in the related art;

FIG. 46 is a waveform diagram illustrating the operation of several components of the power supply circuit shown in FIG. 44 or 45;

FIG. 47 is a graph showing the characteristics of switching frequency and AC.fwdarw.DC power conversion efficiency with respect to the load variation regarding the power supply circuit shown in FIG. 44 or 45; and

FIG. 48 is a graph illustrating a constant voltage controlling characteristic of the power supply circuit shown in FIG. 44 or 45.

DETAILED DESCRIPTION

FIG. 1 shows an example of the configuration of a switching power supply circuit according to a first embodiment of the present invention. Referring to FIG. 1, the power supply circuit is configured such that the basic configuration of the primary side thereof is a combination of a partial voltage resonance circuit with a separated excited current resonance type converter of a half-bridge coupling type.

Further, the power supply circuit of the first embodiment has a configuration ready for a wide range. The circuit operates in response to commercial AC power supplies of both the AC 100 V type and the AC 200 V type. Further, the power supply circuit is ready for a range of variation of the load power Po, for example, from Po=approximately 150 W (100 W or more) to Po=0 W (no load).

Furthermore, the power supply circuit is supposed to be used as a power supply, for example, of a printer apparatus and is configured so as to be ready for a load power Po from 150 W to 0 W.

First, in the power supply circuit shown in FIG. 1, a common mode noise filter is provided for a commercial AC power supply AC and is formed from a pair of filter capacitors CL and a common mode choke coil CMC.

Further, a full-wave rectification circuit including a bridge rectification circuit Di and a smoothing capacitor Ci is connected to the commercial AC power supply AC at the stage following the noise filter.

The full-wave rectification circuit receives the commercial AC power supply AC to perform a full-wave rectification operation such that a rectification smoothed voltage Ei (DC input voltage) is obtained across the smoothing capacitor Ci. The rectification smoothed voltage Ei in this instance has a voltage level equal to that of an AC input voltage VAC.

A current resonance type converter for receiving and switching (interrupting) the DC input voltage includes a switching circuit formed from two switching elements Q1 and Q2 each in the form of a MOS-FET connected in a half-bridge connection as seen in FIG. 1. Damper diodes DD1 and DD2 are connected in parallel between the drain-source of the switching elements Q1 and Q2, respectively. The anode and the cathode of the damper diode DD1 are connected to the source and the drain, respectively, of the switching element Q1. Similarly, the anode and the cathode of the damper diode DD2 are connected to the source and the drain, respectively, of the switching element Q2. The damper diodes DD1 and DD2 are provided as body diodes for the switching elements Q1 and Q2, respectively.

Further, a primary side partial resonance capacitor Cp is connected in parallel between the drain-source of the switching element Q2. The capacitance of the primary side partial resonance capacitor Cp and the leakage inductance L1 of a primary winding N1 cooperatively form a parallel resonance circuit (partial voltage resonance circuit). A partial voltage resonance operation is obtained in which the switching elements Q1 and Q2 exhibit voltage resonance only when the switching elements Q1 and Q2 are turned off.

An oscillation driving circuit 2 is provided to drive the switching elements Q1 and Q2 for switching. The oscillation driving circuit 2 includes an oscillation circuit and a driving circuit and may be formed, for example, using an IC for universal use. The oscillation circuit of the oscillation driving circuit 2 generates an oscillation signal of a required frequency, and the driving circuit makes use of the oscillation signal to generate a switching driving signal used as a gate voltage for driving a MOS-FET for switching. The switching driving signal is applied to the gates of the switching elements Q1 and Q2. Consequently, the switching elements Q1 and Q2 perform a switching operation such that they successively and alternately turn on and off in accordance with a switching frequency corresponding to the period of the switching driving signal.

An insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side.

The insulating converter transformer PIT has the primary winding N1. The primary winding N1 is connected at one end thereof to a node (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 through a series connection with a primary side series resonance capacitor C1 so that the switching outputs are transmitted. The primary winding N1 is connected at the other end thereof to the primary side ground.

The insulating converter transformer PIT has the structure shown in the sectional view of FIG. 2. Referring to FIG. 2, the insulating converter transformer PIT includes an EE type core (EE-shaped core) formed from a pair of E-type cores CR1 and CR2 made of a ferrite material and combined such that magnetic legs thereof are opposed to each other.

The insulating converter transformer PIT further includes a bobbin B made of, for example, a resin material and having such a divisional shape that winding receiving portions on the primary side and the secondary side thereof are independent of each other. The primary winding N1 is wound on one of the winding receiving portions of the bobbin B. Meanwhile, a secondary winding N2 is wound on the other winding receiving portion. The bobbin B on which the primary winding N1 and the secondary winding N2 are wound in this manner is attached to the EE type core (CR1, CR2). Consequently, the primary side winding and the secondary side winding are wound in different winding regions on the inner magnetic leg of the EE type core. Thus, the structure of the insulating converter transformer PIT as a whole is obtained.

A gap G is formed in the manner seen in FIG. 2 in the inner magnetic leg of the EE type core. The gap G in this instance is formed such that the gap length thereof is set, for example, to approximately 2.8 mm so that the coupling coefficient k between the primary side and the secondary side may be, for example, k=0.65 or less, indicating a loose coupling state. Actually, the coupling coefficient k was set to k=0.63. The gap G can be formed by forming the inner magnetic leg of each of the E-type cores CR1 and CR2 shorter than the outer two magnetic legs.

Incidentally, in power supply circuits in the related art which include a current resonance type converter such as the power supply circuit described hereinabove with refere