Title: Wound field synchronous machine control system and method
Abstract: A wound field synchronous machine control system comprises: an auxiliary winding for obtaining auxiliary AC voltage from the wound field synchronous machine; a phase controlled rectifier for rectifying the auxiliary AC voltage and supplying rectified DC voltage to the wound field synchronous machine; and a controller. The controller is configured for using a voltage signal across the auxiliary winding to obtain volt-second values of the auxiliary winding and using the volt-second values for firing angle control of switches of the phase controlled rectifier. Alternatively or additionally, the controller is configured for obtaining airgap flux values of the wound field synchronous machine and using the airgap flux values for firing angle control of switches of the phase controlled rectifier.
Patent Number: 6,870,350 Issued on 03/22/2005 to Garrigan, et al.
| Inventors:
|
Garrigan; Neil Richard (Niskayuna, NY);
Young; Henry Todd (North East, PA)
|
| Assignee:
|
General Electric Company (Niskayuna, NY)
|
| Appl. No.:
|
429545 |
| Filed:
|
May 2, 2003 |
| Current U.S. Class: |
322/28; 322/52; 322/59; 318/700 |
| Intern'l Class: |
H02P 007//36; H02P 011//00; H02P 009//00; H02P 009//40; H02H 007//06 |
| Field of Search: |
322/52,59,89,25,27,28,38,54,26,24,47,36
318/700,716
|
References Cited [Referenced By]
U.S. Patent Documents
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|
| 4161681 | Jul., 1979 | Rathje | 318/783.
|
| 4218729 | Aug., 1980 | Chambers | 363/54.
|
| 4468603 | Aug., 1984 | Vander Meer et al. | 318/779.
|
| 4495449 | Jan., 1985 | Black et al. | 318/60.
|
| 4719361 | Jan., 1988 | Brubaker | 290/45.
|
| 5027285 | Jun., 1991 | McCartney et al. | 324/207.
|
| 5300842 | Apr., 1994 | Lyons et al. | 310/90.
|
| 5334923 | Aug., 1994 | Lorenz et al. | 318/805.
|
| 5652485 | Jul., 1997 | Spiegel et al. | 318/147.
|
| 5729113 | Mar., 1998 | Jansen et al. | 318/799.
|
| 5864217 | Jan., 1999 | Lyons et al. | 318/652.
|
| 5929612 | Jul., 1999 | Eisenhaure et al. | 322/47.
|
| 6011357 | Jan., 2000 | Gradzki et al. | 315/224.
|
| 6081084 | Jun., 2000 | Crecelius | 318/254.
|
| 6396236 | May., 2002 | Luukko | 318/700.
|
| 6417650 | Jul., 2002 | Stefanovic et al. | 322/47.
|
| 6433506 | Aug., 2002 | Pavlov et al. | 318/804.
|
| 6486568 | Nov., 2002 | King et al. | 307/66.
|
Primary Examiner: Schuberg; Darren
Assistant Examiner: Gonzalez; Julio
Attorney, Agent or Firm: Agosti; Ann M., Patnode; Patrick K.
Parent Case Text
This application is a division of application Ser. No. 09/988,129, filed
Nov. 19, 2001 now U.S. Pat. No. 6,586,914, which is hereby incorporated by
reference in its entirety.
Claims
What is claimed is:
1. A flux observer for a wound field synchronous machine coupled to a
direct current load, the flux observer comprising analog electronics,
digital electronics, or a combination of analog and digital electronics
configured receive machine field and direct current load currents from
sensors and to to use the sensed machine field and direct current load
currents to provide a magnetizing flux estimate.
2. The flux observer of claim 1 wherein the flux observer comprises a load
saturation model for receiving the sensed machine field and direct current
load currents and providing the magnetizing flux estimate.
3. The flux observer of claim 1 wherein the flux observer comprises
a load saturation model for receiving the sensed machine field and direct
current load currents and providing a preliminary magnetizing flux
estimate; and
a correction function for receiving a direct axis machine auxiliary current
and providing an auxiliary current correction factor;
a summation element using the preliminary magnetizing flux estimate and the
auxiliary current correction factor to obtain the magnetizing flux
estimate.
4. A method for observing flux in a wound field synchronous machine coupled
to a direct current load, the method comprising sensing machine field and
direct current load currents and using the sensed machine field and direct
current load currents to provide a magnetizing flux estimate.
5. The method of claim 4 wherein using the sensed machine field and direct
current load currents comprises using a load saturation model for
receiving the sensed machine field and direct current load currents and
providing the magnetizing flux estimate.
6. The method of claim 5 wherein sensing occurs while exciting a field
winding of the wound field synchronous machine and while maintaining an
auxiliary winding of the wound field synchronous machine as an open
circuit.
Description
BACKGROUND
The invention relates generally to field regulation of wound field
synchronous machines and more particularly to off-highway vehicle
alternator controls.
Self-propelled traction vehicles such as large off-highway haulage trucks
include electric propulsion systems. A typical propulsion system, such as
described in commonly assigned Black et al., U.S. Pat. No. 4,495,449,
includes an operator-controlled throttle adapted to control the rotational
speed of a prime mover which may, for example, comprise a diesel engine.
An output shaft of the prime mover is drivingly coupled to a rotor of an
alternating current (AC) generator (a wound field synchronous machine
commonly referred to as an alternator) which has a set of three-phase main
windings, an auxiliary (tertiary) winding, and a field winding. The
three-phase, generally sinusoidal, alternating voltage that is generated
in the main windings of the alternator is converted to direct voltage by
means of an uncontrolled full-wave rectifying bridge whose output in turn
is coupled either (a) to at least one armature of a respective adjustable
speed direct current (DC) traction motor or (b) through an inverter to a
respective alternating current (AC) traction motor. The motor rotor is
coupled through suitable speed-reduction gearing to a pair of wheels
located on opposite sides of the vehicle. For an AC traction system, by
controlling the speed of the engine, the excitation of the alternator, and
the inverter torque commands, the vehicle can be propelled (also known as
"motoring") or dynamically retarded (also known as "electric braking") by
the vehicle's motor or motors in either a forward or a reverse direction.
For DC traction motors, during the motoring mode of operation, the motor
will rotate at a speed that depends on both the magnitude of excitation
current in the motor field and the magnitude of the voltage applied to the
armature windings. For AC traction motors, a more complex voltage control
system is typically implemented through one set of armature windings to
control field excitation and torque producing armature current.
The magnitude of the voltage applied to the armature windings is a function
of both the speed at which the alternator is driven and the magnitude of
excitation current in the alternator field. The alternator field
excitation current is supplied by the field winding of the alternator via
a single-phase, full-wave "phase controlled" rectifying bridge. Alternator
field excitation current magnitude depends on the timing of periodic
firing signals that are supplied to the rectifier from a firing angle
control of a controller.
Present implementations for the regulation and control of the field and
output voltage in wound rotor synchronous alternators are subject to
parameter and signal variations. Typically the rectifier that supplies DC
current to the alternator field winding comprises a thyristor such as a
silicon controlled rectifier (SCR) bridge, for example. Thyristor based
(or other phase control based) rectification experiences inherent
non-linear behavior, and thus control presents several control challenges.
An exemplary description of rectifier circuits is provided in JOHANNES
SCHAEFER, RECTIFIER CIRCUITS: THEORY AND DESIGN 1-126 (John Wiley & Sons,
Inc. 1965). Conventional phase-controlled rectifier systems include
techniques based on analog circuitry wherein AC voltage is rectified to
form DC voltage that is applied to the alternator field winding. The
average value of the DC voltage is modulated or controlled by varying the
firing angle of the rectifier bridge. To accomplish the modulation, a ramp
waveform that is synchronous in phase and frequency to the rectified AC
voltage is compared to a small signal reference command signal. The
crossing of the two signals establishes the timing of the turn-on commands
that switch the rectifier bridge.
The analog circuitry of conventional techniques is inherently inflexible to
modifications in that any design changes require hardware changes.
Additionally, the gain of the circuitry is non-linear and highly sensitive
to the operating point of the firing angle, the speed of the alternator,
the level of field excitation, and the load variation. The gain is
additionally sensitive to other variations in the AC voltage such as
distortion due to temperature induced variation. The speed and field
excitation level both change the amplitude of the AC voltage which leads
directly to changes in the amount of voltage applied given a certain
firing angle. In addition, the small signal gain from the reference to the
field voltage is based on a time-averaged value of the discrete pulses of
field voltage. This relation imposes limitations on the outer control
loops in that the bandwidths must be significantly lower (typically on the
order of about ten) than the pulse frequency. The inherent non-linear
transfer function of the time-averaged value imposes further constraints
on the outer control loops. The outer control loops must be stabilized for
all operating points, which means that performance will be compromised to
ensure stability at the worst case operating points. For example, the
outer control loop gains and bandwidths are often set to be sufficiently
low so as to accommodate the least stable operating points of the
rectifier bridge to ensure overall stability.
It would therefore be desirable to have a wound field synchronous machine
control system that is robust to parameter and operating point variations,
insensitive to non-linearities, and readily adaptable to design
modifications.
BRIEF DESCRIPTION
Briefly, in accordance with one embodiment of the present invention, a
wound field synchronous machine control system comprises: an auxiliary
winding for obtaining auxiliary AC voltage from the wound field
synchronous machine; a phase controlled rectifier for rectifying the
auxiliary AC voltage and supplying rectified DC voltage to the wound field
synchronous machine; and a controller for using a voltage signal across
the auxiliary winding to obtain volt-second values of the auxiliary
winding and using the volt-second values for firing angle control of
switches of the phase controlled rectifier.
In accordance with another embodiment of the present invention, a control
system comprises: a sensor for obtaining a voltage signal across a
winding; a phase controlled rectifier for rectifying AC voltage and
supplying rectified DC voltage; and a controller for using the voltage
signal to measure voltage integrals, and using the voltage integrals to
detect zero crossings, obtain volt-second values, and synchronize firing
angle control of switches of the phase controlled rectifier.
In accordance with another embodiment of the present invention, a wound
field synchronous machine control system comprises: an auxiliary winding
for obtaining auxiliary AC voltage from the wound field synchronous
machine; a phase controlled rectifier for rectifying the auxiliary AC
voltage and supplying rectified DC voltage to the wound field synchronous
machine; and a controller for obtaining airgap flux values of the wound
field synchronous machine and using the airgap flux values for firing
angle control of switches of the phase controlled rectifier.
In accordance with another embodiment of the present invention, a flux
observer for a wound field synchronous machine coupled to a direct current
load is configured to use sensed machine field and direct current load
currents to provide a magnetizing flux estimate.
In accordance with another embodiment of the present invention, a wound
field synchronous machine control system comprises: a voltage sensor for
obtaining a voltage signal from the wound field synchronous machine; a
phase controlled rectifier for rectifying AC voltage and supplying
rectified DC voltage to the wound field synchronous machine; a controller
for using the voltage signal to obtain volt-second values, obtaining
airgap flux values of the wound field synchronous machine, and using the
volt-second values and airgap flux values for firing angle control of
switches of the phase controlled rectifier.
In accordance with another embodiment of the present invention, a wound
field synchronous machine control system comprises: AC voltage sensors for
sensing AC phase voltage signals from the machine; a phase controlled
rectifier for rectifying the AC phase voltages and supplying rectified DC
voltage; a DC voltage sensor for sensing a DC voltage signal from the
phase controlled rectifier; and a controller for selectively using the AC
phase voltage signals and the DC voltage signal to estimate a DC load
voltage signal, obtaining a difference between a load voltage command and
the estimated DC load voltage signal, and using the difference for
controlling operation of the synchronous machine.
In accordance with another embodiment of the present invention, a wound
field synchronous machine control system comprises: AC voltage sensors for
sensing AC phase voltage signals from the machine; a phase controlled
rectifier for rectifying the AC phase voltages and supplying rectified DC
voltage to a load; and a controller for using the AC phase voltage signals
to estimate a DC load voltage signal, obtaining a difference between a
load voltage command and the estimated DC load voltage signal, and using
the difference for controlling operation of the synchronous machine.
BRIEF DESCRIPTION OF THE DRAWINGS
These and other features, aspects, and advantages of the present invention
will become better understood when the following detailed description is
read with reference to the accompanying drawings in which like characters
represent like parts throughout the drawings, wherein:
FIG. 1 is a block diagram of a control system in accordance with several
embodiments of the present invention.
FIG. 2 is a graph illustrating volt-seconds on an auxiliary winding with
respect to time.
FIG. 3 is a block diagram illustrating a process for using the auxiliary
winding voltage to control gating signals in accordance with an embodiment
of the present invention.
FIG. 4 is a graph illustrating waveforms applicable to the block diagram of
FIG. 3.
FIG. 5 is a state diagram illustrating states in an example embodiment for
zero crossing detection in accordance with an embodiment of the present
invention.
FIGS. 6-9 are block diagrams of airgap flux estimation models in accordance
with several embodiments of the present invention.
DETAILED DESCRIPTION OF THE INVENTIONS
FIG. 1 is a block diagram of a control system 52 in accordance with several
embodiments of the present invention which may be used individually or in
combination. As discussed above, the control system provides a regulated
DC voltage for powering a load 14 which, in one embodiment, comprises a
traction vehicle's motor drive system. A wound field synchronous machine
12 such as an alternator, for example, is mechanically driven from a prime
mover 10 such as an engine, over a range of speed. The machine field is
electrically excited from a battery source (not shown) during start-up and
electrically self-excited from an auxiliary (tertiary) winding 18 during
normal operation.
A phase-controlled rectifier 20, such as a thyristor bridge, for example,
is used to self-excite a machine field winding 16 by rectifying the AC
auxiliary voltage and applying the resulting DC voltage to the field
winding. In one embodiment, phase-controlled rectifier 20 comprises a
silicon controlled rectifier bridge comprising a plurality of thyristors
43 and 47 having switches 44 and being coupled in parallel to a plurality
of diodes 45. Although auxiliary winding 18 is shown as coupled to
rectifier 20 for purposes of example, such coupling is not required.
An additional rectifier 24, which may comprise a full bridge three-phase
diode output rectifier, for example, converts AC voltage on main stator
windings 22 of the wound field synchronous machine to DC load voltage for
use by load 14. If load 14 includes an AC machine requiring AC voltage,
load 14 may be coupled directly to alternator 12, rectifier 24 may
comprise an AC to AC rectifier, or load 14 may include a DC-AC inverter to
convert the DC voltage from rectifier 24 to AC voltage for the AC machine.
The control system regulates the load voltage by sensing or otherwise
obtaining or calculating appropriate parameters and using a controller 26.
Depending on the application, the load voltage signal may be obtained
using a DC load voltage sensor 28, AC load voltage sensors 29, or a
combination of DC and AC load voltage sensors.
In embodiments where the load can regenerate and reverse bias the rectifier
24, the control system 26 cannot be regulated on the basis of the DC load
voltage (V.sub.ODC) sensed by sensor 28. In these embodiments, a voltage
estimator 35 can be used to estimate a load voltage V.sub.L from the AC
phase-to-phase voltages. In one embodiment, the voltage estimator
comprises a full wave rectifier 31 for rectifying the three phase voltages
and a low pass filter 33 for low pass filtering the rectified voltage and
applying a correction factor based on load current. In a more specific
embodiment, rectifier 31 is designed to compute the maximum instantaneous
difference between any two of the three AC phase voltages. Regardless of
the embodiment selected for DC voltage estimation from AC phase voltages,
the DC voltage estimator can use the DC voltage estimate based on the AC
phase voltages exclusively, switch between the estimate and the sensed DC
voltage as desired, or otherwise use a combination of the two voltage
estimates. In one embodiment, for example, the DC voltage signal is used
except that whenever the DC voltage signal is greater than the AC derived
voltage estimate, the AC derived voltage estimate is used.
The resulting DC load voltage signal (estimate V.sub.L) is supplied to a
summation element 34 along with a load voltage command V.sub.L *.
Regulator 36 uses the difference (error) between V.sub.L * and V.sub.L
from summation element 34 to provide a small signal field voltage command
V.sub.f * (representing a scaled representation of the average value of
the field voltage command) or a volt-second integral command
.DELTA..lambda.* to a firing angle control 38 which controls
phase-controlled rectifier 20 by turning on and off switches 44 and drives
the error to zero through an outer control loop 42. Firing angle control
38 and rectifier 20 are sometimes collectively referred to as a modulator
37. Modulator 37, although shown as a voltage modulator for purposes of
example, may alternately comprise a flux modulator.
Controller 26 may comprise analog, digital, or a combination of analog and
digital electronics. In one embodiment of the present invention, control
algorithms are implemented in controller 26 to enhance regulation of DC
load voltage. In this embodiment, the analog-based phase-controlled
rectifier electronics of conventional control systems are replaced with
analog and digital electronics including a micro-processor with associated
software and programmable logic. Analog electronics, while not required,
remain useful, particularly in the firing angle control 38 for high power
embodiments.
In a more specific embodiment of the present invention, the wound field
synchronous machine 12 control system 52 comprises: auxiliary winding 18
for obtaining auxiliary AC voltage from the wound field synchronous
machine; phase controlled rectifier 20 for rectifying the auxiliary AC
voltage and supplying rectified DC voltage to the wound field synchronous
machine; and controller 26 for using a voltage signal (which preferably
comprises a voltage waveform) across the auxiliary winding to obtain
volt-second values of the auxiliary winding and for using the volt-second
values for firing angle control of switches 44 of the phase controlled
rectifier.
Volt-seconds represent an integration of voltage over time. For example,
one volt-second is equivalent to one volt applied for one second.
Volt-seconds relate to magnetic flux in that a "Weber" is a unit of
magnetic flux whose decrease to zero when linked with a single turn
induces in the turn a voltage with a time integral of one volt-second.
Controller 26 can advantageously use the last available (most recently
obtained) volt-second values for directly calculating the next firing
angle time and thus account for variations in the volt-seconds available
due to distortion, prime mover speed changes, airgap flux changes, and
other variations such as those due to temperature changes. Use of
volt-second calculations results in an accurate modulation function across
the phase-controlled rectifier that is robust to operating point and
parameter variations and that overcomes some difficulties with inherent
non-linearities of conventional ramp comparison control.
To measure the last volt-seconds from auxiliary winding voltage sensor 32
and to synchronize commutation timing of phase-controlled rectifier 20, it
is useful to implement a zero crossing detection algorithm on the AC
auxiliary voltage signal (waveform). In one embodiment, the control system
is configured to use the auxiliary voltage signal to measure voltage
integrals, and use the voltage integrals to detect zero crossings, obtain
the volt-second values, and synchronize the firing angle-control of the
switches. In a more specific embodiment, the voltage integrals are
measured in positive and negative directions (that is, the present and
last volt-seconds). In these embodiments, zero crossings can be detected
in a robust manner with minimal phase delay. In contrast, conventional
zero crossing detection embodiments use simple voltage reference
comparisons that can be subject to noise glitches and distortion that can
cause unwanted miss-fires of switches 44 and subsequent voltage transient
distortion on the DC load voltage.
A more specific embodiment for determining the firing angle from the
previous value of volt-seconds measured on the auxiliary winding is
described as follows with respect to FIGS. 2-5. FIG. 2 is a graph
illustrating volt-seconds on auxiliary winding 18 with respect to time
wherein time t.sub.X represents the time in the present volt-seconds
dividing the present volt-seconds that have passed from the estimated
remaining volt-seconds. FIG. 3 is a block diagram illustrating a process
for using the auxiliary winding voltage V.sub.t to control gating signals
to control firing angles of switches 44, and FIG. 4 is a graph
illustrating pertinent waveforms. Typically the controls for the
embodiment of FIG. 3 are situated within firing angle control 38 of FIG.
1.
The total volt-seconds for each rectified half cycle of the auxiliary
voltage waveform is determined by using an integrator 54 (FIG. 3). A
timing signal (ZCD) from zero cross detection block 56 that is
synchronized with the zero-crossings of the auxiliary waveform is used to
synchronize and reset the volt-second integrals each cycle. More
specifically, the timing signal is synchronized to the auxiliary voltage
waveform zero crossings to reset integrator 54.
At the start of each half cycle, the value of integrator 54
(.lambda..sub.t) is stored in memory 58 as the last volt-second integral
(.lambda..sup.-1). Then integrator 54 is reset to zero and integration of
the next half cycle begins. The commanded integral (.DELTA..lambda.*) of
regulator 36 is then compared at comparator 62 with a signal that is equal
to the last volt-second integral minus the present volt-second integral
(.DELTA..lambda..sub.t =.lambda..sub.t.sup.-1 -.lambda..sub.t as obtained
by summation element 60). When .DELTA..lambda.*equals
.DELTA..lambda..sub.t, the remaining volt-seconds to occur are
substantially equal to the commanded value, and the thyristors are then
gated. In other words, the present volt-seconds .lambda..sub.t are tracked
until is it is estimated based upon the last volt-second integral
.lambda..sup.-1 that the remaining volt-seconds .DELTA..lambda..sub.t will
give the commanded value .DELTA..lambda.*.
As long as variations in the auxiliary waveform are slow or small over the
period of a half cycle, the consecutive half cycle volt-second integrals
will be approximately equal, and the algorithms will work well. Very rapid
variations in the auxiliary waveforms over the period of a half cycle will
introduce errors in the applied volt-second as compared to the command. It
is expected that such errors will typically be small and can be
compensated for by outer control loop 42 voltage regulator 68 (which in
one embodiment, comprises a proportional integral controller for
integrating the error from subtractor 34 and generating a flux command
.lambda.*). Calculation and gating of the thyristors based on a measured
value of the most recent volt-seconds available minimizes the sensitivity
of accuracy to auxiliary waveform distortion and to variations in speed
and flux level. In addition, direct calculation of the firing angle and
gating in this manner ensures that the commanded flux is achieved in no
more than about one half electrical cycle of time.
The zero crossing technique of block 56 may include using hysterisis to
provide immunity to noise and false zero crossing detection. Using
hysterisis, however, involves a tradeoff in that using hysterisis adds
phase delay which can introduce error in the last and present volt-second
calculation.
In another embodiment, zero crossing detection is combined with last
volt-second calculation. In this embodiment, two integrations are used to
calculate the last volt-second, one for each polarity of the signal. When
a potential zero crossing is detected, the opposite polarity integration
begins. If the voltage maintains the new polarity until the integral
reaches a hysteresis threshold, then the new polarity is declared, and the
old polarity integration is reset to zero (ready to start at the next
crossing). If the voltage reverts back to the old polarity before the
hysterisis threshold is reached, it is assumed that the potential zero
crossing was due to noise and not an actual zero crossing, the new
integration is reset to zero (ready for another potential crossing) and
the old polarity integration is continued. During the transition states
before the new polarity is declared, firing of the thyristors is
suppressed. This embodiment is useful because no phase delay or error in
the last volt-second integral is introduced. The only performance
limitation is that switches in rectifier 20 cannot be fired during the
transition state. The integral threshold can be adjusted based on
frequency and amplitude of the auxiliary voltage signal to provide
constant hysterisis angle of the waveform. The integrators for the zero
crossing detector and for the last volt-second do not have to be the same.
In one implementation, for example, the zero crossing detector is
performed in a first microcontroller (not shown) of controller 26 and the
last volt-second integration is done in a Field Programmable Gate Array
(FPGA) (not shown) of controller 26.
As shown in FIG. 5, in one embodiment the auxiliary voltage signal has five
polarity states: 1) Off--system is not enabled to run; 2)
Positive--voltage is positive and firing of a positive thyristor (shown as
thyristor 43 in FIG. 1) is enabled; 3) Maybe negative: voltage is
negative, but voltage integral has not exceeded threshold--thyristor
firing is disabled; 4) Negative--voltage is negative, and firing of a
negative thyristor (shown as thyristor 47 in FIG. 1) is enabled; and 5)
Maybe Positive--voltage is positive, but the integral of the voltage has
not exceeded the threshold to declare positive state-firing is disabled.
In this embodiment, the negative volt-second integration is reset to zero
in the positive state; the positive volt-second integration is reset to
zero in the negative state; the negative volt-second integral is copied
into the last volt-second buffer on the transition from maybe positive to
positive; and the positive volt-second integral is copied in to the last
volt-second buffer on the transition from maybe negative to negative.
Referring again to FIG. 1, in another embodiment of the present invention,
controller 26 of wound field synchronous machine 12 control system 52 is
configured for obtaining airgap flux (magnetizing flux common to all
terminals) values of the wound field synchronous machine and using the
airgap flux values for firing angle control of switches 44 of phase
controlled rectifier 20.
Controlling airgap flux directly as an inner control loop 40 variable (as
compared to conventional techniques of controlling field current) is
advantageous because the DC load voltage V.sub.L is more closely
(mathematically and physically) related to airgap flux than to field
current. The open circuit AC output voltage V.sub.O is directly
proportional to the product of airgap flux and rotational speed (Faraday's
law). The rectified load voltage (V.sub.L) differs from the open circuit
voltage (V.sub.O) due to output impedance (i.e. leakage reactance and
winding resistance), rectification losses and armature reaction. The
effects of output impedance and rectification are small effects. The
armature reaction is due to a changing level of airgap flux resulting from
the magnetizing component of the load current. The armature reaction does
not disturb the relationship of proportionality to airgap flux and can be
compensated for in the estimation of airgap flux level.
Another advantage of controlling airgap flux directly as an inner control
loop 40 variable is that the applied variable of control can have fixed
time duration of field voltage measured in terms of volt-seconds or flux
(because field excitation with voltage over time is substantially
proportional to airgap flux). When the field winding is excited from a
source of voltage, the voltage is applied over fixed controlled intervals
of time such that the variable of control is really volt-seconds or flux.
Together these two properties make the use of an inner flux regulation
control loop advantageous as compared to an inter field current regulation
control loop. Although field current is conveniently measured and
typically the inner control loop variable, the field current has two
limitations. First the field current is magnetically non-linear with flux,
and second the field current is only one of the magnetizing components of
current associated with the airgap flux.
In one embodiment, inner control loop 40 comprises: a flux observer 66 for
receiving current values and estimating a value of feedback flux; a
summation element 64 to subtract an estimated value of feedback flux from
a commanded value of flux; a regulator 36 (modulator); and a controllable
DC voltage source driving the field winding of the machine. In one
embodiment, last volt-second thyristor control is used as described above.
Alternatively, conventional thyristor controls may be used. The field
excitation source may comprise either the auxiliary winding or another
external or internally derived source. The regulation algorithm may
comprise a dead-beat type of control as discussed below or a traditional
type of control such as proportional integral (PI) control.
Overall operation of the inner control loop 40 is as follows. The flux
command .lambda.* (generated by voltage regulator 68) is compared to the
estimated value of airgap flux .lambda. (from flux observer 66) to
generate a flux error which is driven to zero by flux regulator 70. The
flux error is processed in flux regulator 70 according to a flux
regulation algorithm into a voltage (V.sub.f *) or volt-second
(.DELTA..lambda.*) command that is synthesized and applied to the field
terminals through firing angle control 38. Flux regulator 70 may comprise
a conventional proportional integral controller or a "dead-beat" type flux
controller as discussed below.
To obtain the estimated airgap flux .lambda. value, flux observer 66 of
controller 26 can be configured to use current feedback sensor
measurements. Magnetizing current i.sub.m can be expressed as a function
of field current i.sub.f, direct axis stator current i.sub.ds, and direct
axis auxiliary current i.sub.dt :
i.sub.m =i.sub.f +i.sub.ds +i.sub.dt.
Magnetizing flux is a non-linear algebraic function of the magnetizing
current:
.lambda..sub.m =.function.(i.sub.m).
However, using a magnetizing current for flux observation can be
inconvenient because such a technique requires measurement of two phases
of the AC current i.sub.o (to obtain i.sub.ds) and subsequent resolution
of the magnetizing component. This entails the standard d-q synchronous
reference frame coordinate transformations where the three phase AC
variables are transformed into two phase variables of a reference frame
that is synchronous with the machine rotating flux. It is desirable to
avoid these operations to eliminate the required sensors and the
processing algorithms.
In one embodiment, as shown in FIG. 6, field voltage V.sub.f (obtainable,
for example, from sensor 27 of FIG. 1) and field current i.sub.f
(obtainable, for example, from sensor 30 of FIG. 1) are used to estimate
the airgap flux. Per Faraday's Law,
V.sub.f =i.sub.f R.sub.f +d.lambda..sub.f /dt (or .lambda..sub.f
=.intg.v.sub.f -i.sub.f R.sub.f), and
.lambda..sub.m =.lambda..sub.f -i.sub.f L.sub.if,
wherein R.sub.f represents field resistance (which can be estimated and
represented as R), .lambda..sub.f represents field flux, and L.sub.if
represents field leakage inductance. Combining the above two equations
yields:
.lambda..sub.m =.intg.v.sub.f -i.sub.f R-i.sub.f L.sub.if,
as represented in FIG. 6 by subtractors 76 and 78, multipliers 80 and 82,
and integrator 84.
In another embodiment, as shown in FIG. 7, the magnetizing current model
and field voltage and current model are combined (with subtractor 90,
proportional integral controller 92, and adder 94) to form a closed loop
flux observer that provides a hybrid estimate of flux {circumflex
over(.lambda.)}.sub.m that is based on the current model at steady state
and at low frequencies (on an order of magnitude lower than the
fundamental frequency of machine 12 of FIG. 1, for example) and on the
voltage model at higher frequencies. Intermediate frequencies can be a
blend of the two models. The voltage model gives rapid response to dynamic
perturbations in voltage, but lacks steady state stability due to drift of
integrator 84 and lower accuracy in resistance values at low frequencies.
The current model (represented by adder 86) provides steady state and low
frequency stability. Function 88 provides the no load magnetizing flux
estimate from the magnetizing current model.
In some embodiments, such as many off highway vehicle alternator
embodiments, direct axis stator current i.sub.ds and direct axis auxiliary
current i.sub.dt are not readily obtainable. One current measurement that
is typically readily available is DC load current i.sub.L. However, there
is no obvious reflection of DC load current to AC side d-axis magnetizing
current.
Load saturation curves are often plotted for alternators. In such plots the
rectified DC output voltage is measured as a function of DC load current
with field current as a running parameter. In one embodiment, load
saturation type data is recorded with magnetizing flux (rather than output
voltage) as a function of field current and load current. In this
embodiment, to obtain the data, the field winding will be separately
excited from a controlled source, and the auxiliary winding open circuit,
having no current, will provide a good measure of the airgap induced
voltage (which is due to the magnetizing flux). Knowing that the airgap
induced voltage is a derivative of the magnetizing flux, and running at
constant frequency, an RMS (root mean squared) equivalent of the
magnetizing flux can be recorded and calculated from the open circuited
auxiliary voltage. Although the auxiliary magnetizing effect is omitted
from the analysis, the effect is expected to be negligible or to be
compensated for by outer control loop 42 (FIG. 1).
FIG. 8 illustrates an embodiment wherein a current model 96 comprises a
load saturation model 98 that uses field current if and load voltage
i.sub.DC to provide a magnetizing flux estimate
.lambda..sub.m.sub..sub.F.DC which, if desired, can be compensated for at
summation block 100 by subtracting an auxiliary current correction factor
obtained from correction block 102 to obtain a more accurate magnetizing
flux estimate .lambda..sub.m. Current model 96 may be used independently
or, as shown in FIG. 8 for purposes of example, in combination with the
voltage model discussed in FIGS. 6-7.
Load saturation model 98 may comprise a look up table, a fitted curve, or a
mathematical function, for example. Because load saturation curves have
parabolic shapes, the curves can be conveniently represented as parabolic
functions where the DC load current is the independent variable and the
field current is an independent coefficient parameter. In one embodiment,
a linear function is fit for the load current and the load current
x-intercepts I.sub.DC.sub..sub.0 (line of zero flux) and the field
current I.sub.F :
I.sub.DC.sub..sub.0 (I.sub.F)=k.sub.DC.sub..sub.0 I.sub.F,
wherein k.sub.DC.sub..sub.0 represents the slope of a line that best fits
the zero flux points (x-intercepts) of DC current with respect to the
field current. Additionally, a function is fit or a look-up table is
generated to represent the machine's magnetization curve
.lambda..sub.m.sub..sub.0 (I.sub.F) with respect to load current. Then
the relation between field current and load current and modulating flux
combinations can be calculated as:
##EQU1##
Wherein .alpha. represents a load curve shape constant for a respective
field current I.sub.F. In other words, each field current has an
associated magnetizing flux and load current curve with a respective
associated load curve shape constant .alpha.. In one example, the
magnetizing flux and load current curves comprise parabolic curves with
the load current for a given curve equaling the respective load current
x-intercept minus the product of .alpha. and the square of the magnetizing
flux.
The magnetizing flux can be calculated as follows:
##EQU2##
The equation for .lambda..sub.m (I.sub.F) can be rewritten by substituting
the above equations for I.sub.DC.sub..sub.0 (I.sub.F) and a
(I.sub.DC.sub..sub.0 ,.lambda..sub.m.sub..sub.0 ) as:
##EQU3##
and is represented for purposes of example in FIG. 9 as a flux function 106
of a current model 104 (with function 105 provides the no load magnetizing
flux estimate from the field current model). Current model 104 may be used
independently or, as shown in FIG. 9 for purposes of example, in
combination with the voltage model discussed in FIGS. 6-7.
Referring again to FIG. 1, in one embodiment, the flux regulation algorithm
of flux regulator 70 is implemented using dead-beat flux control. This
method of control directly uses the last volt-second based calculation of
firing angles. Dead-beat control is a terminology used in discrete time
control systems wherein the output of a regulated system is controlled
such that the commanded value is achieved in a single time step of the
discrete time system. In other words the command is "dead-on" in one time
"beat" of the controller. In this embodiment, the flux is controlled such
that the command is approximately achieved in one half of an electrical
cycle of the machine. The discrete time constraint here of one half
electrical cycle stems from the available conduction interval of the phase
controlled rectifier excited from the auxiliary winding. The discrete time
interval varies with machine speed.
More specifically, in this embodiment of the present invention, controller
26 of wound field synchronous machine 12 control system 52 is configured
for using the voltage signal across the auxiliary winding to obtain
volt-second values of the auxiliary winding, obtaining airgap flux values,
and using the volt-second values and the airgap flux values for firing
angle control of switches 44 of the phase controlled rectifier.
In the dead-beat flux control, the flux error, being in the units of and
equivalent to volt-seconds, is used to directly calculate the firing angle
as described in the last volt-second thyristor control description.
Alternatively a conservative design may use a scaled (percentage of unity)
flux error to determine the firing angle. It is useful for the firing
angle to be selected such that the volt-seconds applied over the
conduction interval exactly equal the flux error such that at the end of
the half cycle the commanded flux is achieved.
The main advantages of the dead-beat flux control is the direct calculation
algorithm and simplicity of implementation and the rapid and stable
response behavior. An alternative approach based on time-averaged control
would employ a regulator that would regulate the value of flux measured
and averaged over many cycles of the discrete time interval. Time averaged
control is widely understood in pulse width modulation control and
thyristor control. A limitation of such control methods is that the time
averaged nature inherently requires many cycles and dictates limited
response times. Instabilities occur when the bandwidths are pushed higher
and approach the discrete time constraint. The dead-beat control pushes
the response time to the fastest possible limit and simultaneously ensures
stability with direct calculation of modulation level.
In another embodiment of the present invention, state feedback control 74
is used to decouple load current (I.sub.L) and engine speed
(.omega..sub.e). Changes in the operating point such as in electrical load
or in speed result in changes in the output voltage representing undesired
disturbances. Closed loop feedback control in outer control loop 42
ultimately modifies the flux level to compensate for such disturbances and
to maintain the output voltage at the commanded reference value. Such
control action takes time to occur as the signals circulate through the
entire feedback control system and is subject to the control loop dynamic
response.
It would be desirable to bypass outer control loop 42 and more quickly
translate the required flux changes through inner control loop 40 to
decouple the disturbances. Any residual error not compensated via the
inner control loop can then be compensated with the outer control loop.
However it can be shown to very effectively minimize the disturbance
effects of variations in load and speed on the output voltage.
In one embodiment, for example, state feedback control 74 receives values
of engine speed and load current and uses an analytical model or a lookup
table to determine what change in flux is needed to compensate for change
in speed and load current and to generate an appropriate state feedback
decoupling flux signal .lambda..sub.sfd to be included at summation
element 64.
In another embodiment of the present invention, the engine speed is
alternatively or additionally fed directly back into voltage regulator 68,
and voltage regulator 68 uses the engine speed, in combination with the
error from 34 to generate the flux command .lambda.*. This can be
accomplished by using the engine speed to adjust a gain schedule of the
voltage regulator.
In another embodiment of the present invention, command feedforward control
72 can be used to track variations in the load voltage command V.sub.L *.
Normal operation of the system may require that the output voltage be
changed according to some predetermined operating conditions such as speed
or mode of operation. During these times, the load voltage command V.sub.L
* is changed. It is desired to have the load voltage V.sub.L track the
command voltage as accurately as possible.
Without feedforward compensation, voltage regulator will detect an error as
the command is changed and will drive outer control loop 42 in such a
fashion as to correct the error and regain tracking of the load voltage
with the load voltage command. This response is subject to the dynamics of
the outer control loop which again may be somewhat slow. A faster and
preferred response can be achieved if the required flux change necessary
to accommodate the changing command is predetermined from the command and
fed forward to the inner control loop at summation element 64. In this
case the inner control loop is much faster and can more rapidly respond to
the command without waiting for outer control loop. Again compensation of
residual error can be accomplished by the outer control loop. The result
is an improved response to variations in the commanded reference value. In
one embodiment, for example, feedforward control 72 receives the load
voltage command and uses an analytical model or a lookup table to
determine what change in flux is needed to compensate for any changes in
the load voltage command and to generate an appropriate feedforward flux
command .lambda..sub.ff * to be included at summation element 64.
While only certain features of the invention have been illustrated and
described herein, many modifications and changes will occur to those
skilled in the art. It is, therefore, to be understood that the appended
claims are intended to cover all such modifications and changes as fall
within the true spirit of the invention.
*
